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United States Patent |
6,215,359
|
Peckham
,   et al.
|
April 10, 2001
|
Impedance matching for a dual band power amplifier
Abstract
An exciter matching circuit (125), interstage matching circuit (134), and
harmonic filter matching circuit (140) match impedances at the input to a
two-stage power amplifier (130), between the first stage (132) and the
second stage (136) of the power amplifier (130), and at the output of the
power amplifier (130) for more than one frequency band of interest. In a
GSM/DCS dual band radiotelephone (101), the matching circuits (124, 134,
140) provide low return loss at 900 MHz when the dual band transmitter
(110) is operating in the GSM mode. The harmonic filter matching circuit
(140) also filters out signals at 1800 MHz, 2700 MHz, and high order
harmonics. When the dual band transmitter (110) is in DCS mode, however,
the matching circuits (124, 134, 140) provide a low return loss at 1800
MHz and filter out signals at 2700 MHz and harmonics of 1800 MHz.
Inventors:
|
Peckham; David Sutherland (Barrington Hills, IL);
Black; Gregory Redmond (Vernon Hills, IL)
|
Assignee:
|
Motorola, Inc. (Schaumburg, IL)
|
Appl. No.:
|
563721 |
Filed:
|
May 1, 2000 |
Current U.S. Class: |
330/302; 330/145; 330/310 |
Intern'l Class: |
H03F 003/191 |
Field of Search: |
330/51,150,302,306,310,144,145
|
References Cited
U.S. Patent Documents
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|
4959873 | Sep., 1990 | Flynn et al. | 455/303.
|
5060294 | Oct., 1991 | Schwent et al. | 455/93.
|
5361403 | Nov., 1994 | Dent | 455/74.
|
5392463 | Feb., 1995 | Yamada | 330/284.
|
5406224 | Apr., 1995 | Mikami et al. | 330/277.
|
5423074 | Jun., 1995 | Dent | 455/74.
|
5432473 | Jul., 1995 | Mattila et al. | 330/133.
|
5455968 | Oct., 1995 | Pham | 330/285.
|
5530923 | Jun., 1996 | Heinonen et al. | 330/51.
|
5532646 | Jul., 1996 | Aihara | 330/279.
|
5748042 | May., 1998 | Norris et al. | 330/277.
|
5774017 | Jun., 1998 | Adar | 330/51.
|
6043721 | Mar., 2000 | Nagode et al. | 330/51.
|
6078794 | Jun., 2000 | Peckham et al. | 330/306.
|
6133793 | Oct., 2000 | Lau et al. | 330/302.
|
Foreign Patent Documents |
0 459 440 B1 | Sep., 1996 | EP.
| |
2 236 028 | Mar., 1991 | GB.
| |
2 260 871 | Apr., 1993 | GB.
| |
Primary Examiner: Mottola; Steven J.
Attorney, Agent or Firm: Chen; Sylvia, Bowler II; Roland K.
Parent Case Text
CROSS REFERENCE TO RELATED APPLICATION
This patent application is a division of U.S. patent application Ser. No.
08/802,831 filed Feb. 19, 1997 by U.S. Pat. No. 6,078,794 David S. Peckham
et al. and entitled "Impedance Matching for a Dual-Band Power Amplifier."
This related application is hereby incorporated by reference herein in its
entirety, and priority thereto for common subject matter is hereby
claimed.
Claims
We claim:
1. A two-stage power amplifier for a dual band constant envelope modulation
communication device selectively operable in a first frequency band and a
second frequency band comprising:
a first stage;
a second stage;
a first capacitance coupled between the first stage and the second stage;
an inductance coupled between a first stage output and a voltage source;
a second capacitance coupled between the voltage source and ground;
a third capacitance;
a diode coupled between the third capacitance and ground; and
a resistor coupled between the diode and a band selection voltage.
2. A two-stage power amplifier according to claim 1, wherein the first
stage is a first saturation mode transistor and the second stage is a
second saturation mode transistor.
3. A two-stage power amplifier according to claim 2, wherein the first
transistor is a metal semiconductor field-effect transistor and the second
transistor is a metal semiconductor field-effect transistor.
4. A dual band transmitter for operation in a saturated mode at either a
first frequency band or a second frequency band, comprising:
a first saturation mode amplifier stage having a first stage output
impedance;
a second saturation mode amplifier stage having a second stage input
impedance;
a variable impedance interstage matching circuit interconnecting the first
stage and the second saturation mode amplifier stages,
the interstage matching circuit having a first impedance configuration when
the first frequency band is selected, the interstage matching circuit
having a second impedance configuration when the second frequency band is
selected,
whereby the interstage matching circuit matches the first stage output
impedance in either the first frequency band or the second frequency band
to the second stage input impedance.
5. A dual band transmitter according to claim 4 wherein the first stage
further comprises:
a first stage input impedance,
wherein the first stage has power efficient operation in a saturated mode
at either the first frequency band or the second frequency band.
6. A dual band transmitter according to claim 4 wherein the second stage
further comprises:
a second stage input impedance,
wherein the second stage has power efficient operation in a saturated mode
at either the first frequency band or the second frequency band.
7. A dual band transmitter according to claim 4 wherein the interstage
matching circuit comprises:
a first capacitance, coupled between the first stage and the second stage;
an inductance, coupled between a first stage output and a voltage source;
a second capacitance, coupled between the voltage source and ground; and
a switch, for connecting a third capacitance between the voltage source and
ground when the first frequency band is selected and disconnecting the
third capacitance from the voltage source when the second frequency band
is selected.
8. A dual band transmitter according to claim 4 wherein the interstage
matching circuit comprises:
a first capacitance coupled between the first saturation mode amplifier
stage and the second saturation mode amplifier stage;
an inductance coupled between an output of the first saturation mode
amplifier stage and a voltage source;
a second capacitance coupled between the voltage source and ground;
a third capacitance coupled between the voltage source and ground and
a switch in series with the third capacitance, the switch having one
configuration connecting the third capacitance between the voltage source
and ground when the first frequency band is selected, the switch having
another configuration disconnecting the third capacitance from between the
voltage source and ground when the second frequency band is selected.
9. A dual band transmitter according to claim 8, the switch comprises a
diode coupled between the third capacitance and ground, a frequency mode
selection voltage source, and a resistor interconnecting the frequency
mode selection voltage source, the third capacitance and the diode.
10. A multi-stage power amplifier selectively operable in one of at least
two different frequency bands, comprising:
a first amplifier stage;
a second amplifier stage;
a variable impedance circuit interconnecting the first and second amplifier
stages, the variable impedance circuit having a switch capacitor coupled
to ground,
a diode coupled between the switch capacitor of the variable impedance
circuit and ground;
a frequency mode selection voltage source;
a resistor coupled between the frequency mode selection voltage source, the
switch capacitor of the variable impedance circuit and the diode.
11. The multi-stage power amplifier of claim 10, the variable impedance
comprises an interconnection capacitor interconnecting the first and
second amplifiers stages, an inductor coupling an output of the first
amplifier stage to a voltage source, a ground capacitor coupled between
the voltage source and ground, the switch capacitor coupled to the voltage
source.
Description
FIELD OF THE INVENTION
This invention relates generally to dual band communication systems, and
more particularly to impedance matching circuits for a power amplifier in
a dual band transmitter.
BACKGROUND OF THE INVENTION
A dual mode transmitter can operate using two different systems. For
example, an AM/FM dual mode transmitter can transmit both amplitude
modulated and frequency modulated signals. For radiotelephones, a dual
band transmitter can operate using two different cellular telephone
systems. For example, a dual band GSM/DCS radiotelephone can use the
Global System for Mobile Communications (GSM), which operates at 900 MHz,
and the Digital Communications System (DCS), which is similar to GSM
except that it operates at 1800 MHz.
In any radiotelephone, the power amplifier at the final stage of the
transmitter should be matched to the impedance of the antenna.
Additionally, harmonics of the transmitted frequency band should be
suppressed to reduce interference with other communication systems
operating at the harmonic frequencies. With a GSM/DCS dual band
transmitter, it is difficult to suppress the first (1800) MHz harmonic
during 900 MHz GSM transmissions and yet pass the 1800 MHz signal during
DCS transmissions. Also the output impedance of a radiotelephone power
amplifier should be matched to the antenna so that the impedance at the
output of the amplifier is at the optimum impedance for power efficient
amplification.
Thus, there is a need for a dual band power amplifier that can suppress
harmonic frequencies during a first mode of transmission and also properly
pass signals during a second mode of transmission, even when the signals
of the second transmission are at or near a harmonic frequency of the
first mode of transmission. There is also a need for a dual mode power
amplifier with a limited number of parts and a low current drain.
BRIEF DESCRIPTION OF THE DRAWINGS
FIG. 1 shows a diagram of a communication system having matching circuits
according to a preferred embodiment.
FIG. 2 shows a diagram of the exciter matching circuit according to the
preferred embodiment.
FIG. 3 shows a diagram of the two-stage power amplifier according to the
preferred embodiment.
FIG. 4 shows a diagram of the harmonic filter matching circuit according to
the preferred embodiment.
FIG. 5 shows a graph of a return loss signal and an attenuation signal at
the output of the harmonic filter matching circuit in GSM mode according
to the preferred embodiment.
FIG. 6 shows a graph of a return loss signal and an attenuation signal at
the output of the harmonic filter matching circuit in DCS mode according
to the preferred embodiment.
DETAILED DESCRIPTION OF THE PREFERRED EMBODIMENT
Three matching circuits enable a modulator, power amplifier, and antenna of
a radiotelephone transmitter to efficiently amplify and transmit signals
at more than one frequency band while suppressing first, second, and
higher order harmonics. An exciter matching circuit matches the impedance
at the output of the modulator to the impedance at the input of the power
amplifier for both modes of a dual band transmitter. An interstage
matching circuit has a switch to match impedances between a first stage
and a second stage of a power amplifier during different bands of
operation. Finally, a harmonic filter matching circuit uses a switch to
match impedances and adjust the filter pass band of a combined filter and
matching circuit during different modes of operation.
FIG. 1 shows a diagram of a communication system 100 having matching
circuits 125, 134, 140 according to a preferred embodiment. The
communication system 100 shown is a cellular communication system with a
handset radiotelephone 101 and a base station transceiver 190, however, a
different communication system could be substituted, such as a
modulator/demodulator (MODEM), a paging system, or a two-way radio system.
The radiotelephone 101 is a dual band GSM/DCS radiotelephone, however,
other transmission modes with constant envelope modulation schemes can be
substituted for either the GSM mode, the DCS mode, or both modes. Other
constant envelope modulation communication systems include Advanced Mobile
Phone Service (AMPS) and ETACS (European Total Access Cellular System),
which use frequency modulation (FM), and Personal Communication System
(PCS) 1900 which uses Gaussian Minimum Shift Keying (GMSK) as does GSM and
DCS. Transmission modes may also be added to create a tri-mode or
quad-mode radiotelephone.
The radiotelephone 101 includes a microphone 105 for picking up audio
signals. In a dual band transmitter 110, the audio signals are coded by a
speech coder 115 and sent to a modulator 120. Depending on the mode in
use, the modulator 120 mixes the coded signals to 900 MHz in the case of
GSM or 1800 MHz in the case of DCS. An exciter matching circuit 125
includes a bipolar junction transistor (BJT) and matches the approximately
50.OMEGA. impedance at the BJT output to the approximately 7.OMEGA.
impedance at the power amplifier 130 input for the frequency bands of
interest, which is either at 900 MHz or 1800 MHz depending on the mode in
use. Power amplifier 130 is preferably a gallium arsenide (GaAs)
field-effect transistor (FET) two-stage amplifier with a first stage 132
and a second stage 136. Other device types, however, such as silicon BJTs
or silicon FETs could be substituted for the GaAs FETs. Between the two
stages is an interstage matching circuit 134 that optimizes the impedance
matching at either 900 MHz or 1800 MHz depending on the mode in use. At
the output of the power amplifier 130, which has an impedance of
approximately 8-10.OMEGA. and sometimes varies depending on the
transmitter mode in use, a harmonic filter matching circuit 140 matches
the outgoing signal to the approximately 50.OMEGA. antenna 155 at the
frequency band of interest and filters out first, second, and higher order
harmonics of the signal. The matched impedances presented to the power
amplifier input and the power amplifier output by the exciter matching
circuit 125 and the harmonic filter matching circuit 140 determine the
efficiency of the power amplifier.
The transmitted signal is received by a complementary transceiver 190, such
as a GSM cellular base station, through an antenna 195. A DCS base station
is also compatible with the GSM/DCS radiotelephone 101, and other
transceivers would be compatible with PCS, AMPS, or ETACS dual mode
radiotelephones. Signals from the base station transceiver 190 are
transmitted from the antenna 195 of the base station and received by the
antenna 155 of the radiotelephone 101. A duplexer 150 in the
radiotelephone 101 controls whether the antenna 155 is transmitting or
receiving signals. Received signals are sent through the duplexer 150 to
receiver 160. In the receiver 160, a radio frequency (RF) receiver 165
prepares the signal for demodulation, a demodulator 170 demodulates the
signal, and a speech decoder 175 decodes the demodulated signal to an
audio format for reproduction on speaker 180.
FIG. 2 shows a diagram of the exciter matching circuit 125 according to the
preferred embodiment. When a GSM signal at 900 MHz emerges from the
modulator 120 (shown in FIG. 1), certain components of the exciter
matching circuit 125 dominate the impedance response to promote a match to
the power amplifier (shown in FIG. 1) at 900 MHz while rejecting other
frequencies. Likewise, when a DCS signal at 1800 MHz comes from the
modulator 120, different components dominate the impedance response of the
exciter matching circuit 125 to create a good match at 1800 MHz while
creating a poor match at other frequencies.
The modulator 120 is isolated from the power amplifier 130 (shown in FIG.
1) using a resistance buffer with resistor 205 and resistor 207. A 1 pF
capacitance 215 is also connected from ground to the base of a BJT 210.
The BJT 210 is used to amplify and transform the impedance of a modulated
signal before the signal enters the power amplifier 130 (shown in FIG. 1).
The output of the BJT is at approximately 50.OMEGA.. A quarter-wave
transmission line 220 is connected from the collector of the BJT 210 to a
constant voltage source V.sub.B2. This transmission line 220 acts as an
inductor when the modulated signal is at 900 MHz and acts as an open
circuit when the modulated signal is at 1800 MHz. A 68 pF capacitance 225
is connected between the voltage source V.sub.B2 and ground, and a
resistor 227 is parallel to the transmission line 220. The resistor 227
stabilizes the BJT by providing a resistive termination when the
transmission line 220 acts as an open circuit. A 4.7 pF capacitance 230 is
also connected to the collector of the BJT 210, which functions as a
direct current (DC) blocking element and as an impedance transforming
element at 900 MHz. Two transmission lines 240, 250 connect the signal
from the capacitance 230 to the output of the exciter matching circuit
125, which connects to the power amplifier 130 (shown in FIG. 1). Between
the two transmission lines 240, 250 is a 1.5 pF capacitance 245 to ground.
During operation, when a 900 MHz GSM modulated signal enters the input to
the exciter matching circuit 125, the inductance of the transmission line
220 and capacitance 230 dominate the impedance of the exciter matching
circuit 125 to create a good match at 900 MHz at approximately 7.OMEGA.
input impedance of the power amplifier 130 (shown in FIG. 1). The other
elements in the exciter matching circuit 125 have a negligible effect on
the impedance at the 1800 MHz frequency band. In other words, the
inductance of the transmission line 220 and the capacitance 230 act as a
high pass filter that also transforms lower frequency signals.
When an 1800 MHz DCS modulated signal enters the exciter matching circuit
125, the transmission line 220 is open and the inductance of transmission
lines 240, 250 and the capacitance 245 dominate the impedance of the
exciter matching circuit 125 to create a good match at 1800 MHz to the
approximately 7.OMEGA. input impedance of the power amplifier 130 (shown
in FIG. 1). In this case, the transmission line 220 and capacitance 230
have a negligible effect on the impedance at the 900 MHz frequency band.
The inductance of the transmission lines 240, 250 and the capacitance 245
act as a low pass filter that also transforms higher frequency signals.
FIG. 3 shows a diagram of the two-stage power amplifier 130 according to
the preferred embodiment. An interstage matching circuit 134 matches the
impedances between the first stage 132 and the second stage 136 of the
two-stage power amplifier 130. The interstage matching circuit 134
optimizes the impedances at 900 MHz or 1800 MHz depending on which
transmission mode is in use.
Two metal semiconductor field-effect transistors (MESFETs) are used as
power amplifier stages 132, 136 in the power amplifier 130. Alternatives
to the MESFETs include silicon BJTs, silicon MOSFETs, and heterojunction
bipolar transistors (HBTs). Between the two stages 132, 136 is a 15 pF
capacitance 325, and at the source of the first stage 132 is a small 3 nH
inductance 335 which is connected to a voltage source V.sub.B3. The two
stages 132, 136, the inductance 335, and the capacitance 325 are
integrated onto a chip 310. Outside of the chip 310, a 2.7 pF capacitance
340 is connected between the inductance 335 and the voltage source
V.sub.B3. A 1000 pF capacitance 350 is also connected to the voltage
source V.sub.B3 with a diode 370 connected from the capacitance 350 to
ground. A 1.5 k.OMEGA. resistor 360 with an input node 365 is connected
between the capacitance 350 and the diode 370.
When a voltage source is connected to the input node 365, the diode 370
turns on and the 1000 pF capacitance 350 dominates the impedance of the
interstage matching circuit 134. The capacitance values are calculated so
that 900 MHz GSM signals from the first stage 132 of the power amplifier
130 are matched to the second stage of the power amplifier 130 (shown in
FIG. 1) when the input node 365 is connected to a 2.7 V positive voltage
source. When a zero, negative, or floating voltage source is connected to
the input node 365, the 2.7 pF capacitance 340 and the 3 nH inductance 335
and the capacitance 350 dominate the impedance of the interstage matching
circuit 134 which then matches 1800 MHz DCS signals to the second stage
136 of the power amplifier 130 (shown in FIG. 1). Thus, the voltage source
applied to node 365 is a GSM/DCS mode selection voltage. Voltage is
applied to node 365 when the radiotelephone 101 is in GSM mode, and
voltage is not applied to node 365 when the radiotelephone 101 is in DCS
mode.
FIG. 4 shows a diagram of the harmonic filter matching circuit 140
according to the preferred embodiment. The harmonic filter matching
circuit 140 uses both impedance matching and low pass filtering to pass
900 MHz signals and suppress 1800 MHz, 2700 MHz, 3600 MHz, and higher
order harmonics during GSM mode transmissions while passing 1800 MHz
signals and suppressing 2700 MHz signals and 3600 MHz and higher order
harmonics during DCS transmissions.
The output of the power amplifier 130 (shown in FIG. 1) is connected
through a first transmission line 410 to a voltage source V.sub.B4.
Transmission line 410 is preferably a half-wave transmission line at 2700
MHz. A 100 pF capacitance 412 is also connected to the voltage source
V.sub.B4.
A set of transmission lines 420, 430, 440, 450 is connected in series to
the output of the power amplifier 130. At the ends of each transmission
line is a connection from an approximately 3 pF capacitance 422, 442, 452,
482 through a diode 415, 425, 435, 445 to ground. The capacitance of each
diode 415, 425, 435, 445 when the diode is off adds a fixed parallel
capacitance to the switched capacitances 422, 442, 452, 483. An additional
1.8 pF capacitance 432 is connected in parallel to the first capacitance
422 and diode 415 pair.
This structure can be described as a cascade of four low-pass matching
sections. The reactance of the first three sections, which include
transmission lines 420, 430, 440, are switchable using diodes 415, 425,
435, 435. Between each capacitance and diode pair is a 1.5 k.OMEGA.
resistor 416, 426, 436, 446 connected to node 465, which controls the
switching of the first three sections. A 100 pF capacitance 434 connects
the node 465 and ground. Additional 1 pF or less capacitances 462, 472,
492, 414, 424, provide attenuation for the 2700 MHz, 3600 MHz and high
order harmonics of the 900 MHz GSM and 1800 MHz DCS signals. The reactance
of the final section, which includes transmission line 450, is fixed. This
final section suppresses 3600 MHz harmonics generated by the diodes 415,
425, 435, 435 when they are off.
When a 2.7 V positive voltage source is applied to node 465, diodes 415,
425, 435, 445 turn on, and the approximately 3 pF capacitances 422, 442,
452, 482 and the inherent inductance in the diodes 415, 425, 435, 445
filter out 1800 MHz signals. Thus, the GSM/DCS mode selection voltage used
for the interstage matching circuit 134 (shown in FIG. 3) can also be used
to control the operation of the harmonic filter matching circuit 140.
Positive voltage is applied to node 465 when the radiotelephone 101 is in
GSM mode, and negative, zero, or floating voltage is applied to node 465
when the radiotelephone 101 is in DCS mode. The operation of the harmonic
filter matching circuit 140 provides impedance matching at 900 MHz when
the GSM mode is selected via node 465 with signal attenuation at the
harmonic frequencies of 1800 MHz, 2700 MHz, and 3600 MHz as well as other
high order harmonic frequencies. When the DCS mode is selected, however,
the harmonic filter matches at 1800 MHz and provides signal attenuation
starting at 2700 MHz as well as 3600 MHz and higher order harmonics.
FIG. 5 shows a graph of a return loss signal 540 and an attenuation signal
550 at the output of the harmonic filter matching circuit 140 (shown in
FIG. 1) in GSM mode according to the preferred embodiment. The X-axis 510
of the graph measures frequency in MHz while the Y-axis 520 of the graph
measures attenuation in dB. The return loss signal 540 has a significant
lowering in return loss signal at 900 MHz, which indicates a good
impedance match at the 900 MHz GSM frequency band. Also, at 900 MHz, the
attenuation signal 550 is close to 0 dB, which passes the 900 MHz signal
at full power. Meanwhile, at 1800 MHz, 2700 MHz, and 3600 MHz, the
attenuation signal 550 lowers to dampen harmonics of the 900 MHz signal.
FIG. 6 shows a graph of a return loss signal 640 and an attenuation signal
650 at the output of the harmonic filter matching circuit 140 (shown in
FIG. 1) in DCS mode according to the preferred embodiment. The X-axis 610
of the graph measures frequency in MHz while the Y-axis 620 measures
attenuation in dB. The return loss signal 640 has a significant lowering
in return loss signal at 1800 MHz, which indicates a good impedance match
at the 1800 MHz DCS frequency band. Also, at 1800 MHz, the attenuation
signal 650 is close to 0 dB, which is very different than the attenuation
signal characteristic for the harmonic filter matching circuit when it is
in the GSM mode. The attenuation signal 650 still lowers at 2700 MHz and
3600 MHz to dampen harmonics of the 1800 MHz signal.
Depending on the systems used in the dual mode radiotelephone 101,
component values of the exciter matching circuit 125, the interstage
matching circuit 134, and the harmonic filter matching circuit 140 can be
adjusted to match only at the frequency bands of interest. Also,
transmission lines within the three matching circuits can be replaced with
inductances to reduce size or to promote fabrication onto an integrated
circuit.
The exciter matching circuit uses impedance characteristics to promote
matching of the modulator output and the power amplifier input of a dual
mode transmitter at more than one frequency band of interest. The matching
characteristics within the exciter matching circuit change depending on
the frequency band of the input signal. The interstage matching circuit
134 uses a switch to add components, which varies the matching
characteristic of the interstage matching circuit between the first stage
and the second stage of a two-stage power amplifier depending on the mode
used by the dual mode transmitter. The harmonic filter matching circuit
140 also uses switches to add components to vary the matching
characteristic and the filter characteristic of the harmonic filter
matching circuit between the output of the power amplifier and the input
of the antenna depending on the mode used by the dual mode transmitter.
Thus, the three matching circuits use very few additional components to
provide matching at more than one frequency band of interest and filter
out undesired harmonics for dual mode transmitters dependent upon the mode
in use. While specific components and functions of the impedance matching
for a dual band power amplifier are described above, fewer or additional
functions could be employed by one skilled in the art within the true
spirit and scope of the present invention. The invention should be limited
only by the appended claims.
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