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United States Patent |
6,049,201
|
Feldtkeller
|
April 11, 2000
|
Circuit configuration for a charge pump and voltage regulator circuit
having a circuit configuration of this type
Abstract
A circuit configuration for a charge pump includes a first series circuit
of n integrators and a second series circuit of n further integrators that
are connected up to one another via feedback paths. The individual
integrators may be configured in the form of controllable integrators
which, when used in a voltage regulator circuit, are set depending on the
output signal of a regulating amplifier. In particular, the frequency of
the charge pump can be adapted in a load-dependent manner with the aid of
the controllable integrators.
Inventors:
|
Feldtkeller; Martin (Munchen, DE)
|
Assignee:
|
Siemens Aktiengesellschaft (Munich, DE)
|
Appl. No.:
|
299366 |
Filed:
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April 26, 1999 |
Foreign Application Priority Data
| Apr 24, 1998[DE] | 198 18 465 |
Current U.S. Class: |
323/288; 323/284; 363/60 |
Intern'l Class: |
G05F 001/40 |
Field of Search: |
323/282,280,288,289,284
363/59,60
|
References Cited
U.S. Patent Documents
5654628 | Aug., 1997 | Feldtkeller | 323/282.
|
5880579 | Mar., 1999 | Wei et al. | 323/282.
|
Foreign Patent Documents |
0 633 662 A1 | Jan., 1995 | EP.
| |
Other References
"Efficiency Improvement in Charge Pump Circuits" (Wang et al.), IEEE
Journal of Solid State Circuits, vol. 32, No. 6, Jun. 1997, pp. 852-860.
|
Primary Examiner: Riley; Shawn
Attorney, Agent or Firm: Lerner; Herbert L., Greenberg; Laurence A., Stemer; Werner H.
Claims
I claim:
1. A circuit configuration for a charge pump, comprising:
a first series circuit of n integrators having a first integrator and a
last integrator generating an output signal;
a first feedback path and a second feedback path;
one of said integrators of said first series circuit generating an output
signal and receiving, as an input signal, a summation signal formed from
said output signal of said one of said integrators of said first series
circuit fed back via said first feedback path, a second input signal and a
third input signal;
a second series circuit of n further integrators having a first integrator
and a last integrator generating an output signal;
a further first feedback path and a further second feedback path;
one of said integrators of said second series circuit generating an output
signal and receiving, as an input signal, a summation signal formed from
said output signal of said one of said integrators of said second series
circuit fed back via said further first feedback path, a second input
signal and a third input signal;
said output signal of said one of said integrators of said second series
circuit fed back via said second feedback path forming said second input
signal of said one of said integrators of said first series circuit;
said output signal of said one of said integrators of said first series
circuit fed back via said further second feedback path forming said second
input signal of said one of said integrators of said second series
circuit;
said output signal of said last integrator of said second series circuit
forming said third input signal of said one of said integrators
corresponding to said first integrator of said first series circuit;
said output signal of said last integrator of said first series circuit
supplying said third input signal of said one of said integrators
corresponding to said first integrator of said second series circuit;
said third input signal for remaining integrators of said first series
circuit supplied by an output signal of a preceding integrator of said
integrators of said first series circuit;
said third input signal for remaining integrators of said second series
circuit supplied by an output signal of a preceding integrator of said
integrators of said second series circuit; and
said integrators of said first series circuit and said second series
circuit having outputs that can be picked off outputting output signals
phase-shifted with respect to one another.
2. The circuit configuration according to claim 1, wherein said first
feedback path and said further first feedback path are assigned a common
first gain factor K1, and said second feedback path and said further
second feedback path are assigned a common second gain factor K2, the
common first gain factor K1 and the common second gain factor K2 being
chosen such that they fulfill the following equations: K2>K1, K2-K1<1 and
K1+K2>1.
3. The circuit configuration according to claim 1, wherein n=2, and said
output signals phase-shifted with respect to one another of said
integrators of said first series circuit and said second series circuit
being four substantially sinusoidal output signals phase-shifted in each
case by 90.degree. with respect to one another and can be picked off at
said outputs of said integrators of said first series circuit and said
second series circuit.
4. The circuit configuration according to claim 1, wherein said integrators
of said first series circuit and said second series circuit are
controllable with regard to their transfer characteristic, current
consumption and maximum switching frequency.
5. The circuit configuration according to claim 1, wherein each of said
integrators is an inverter controllable by a plurality of control signals.
6. The circuit configuration according to claim 5, wherein each of said
integrators has controllable resistors and an output current set by said
controllable resistors.
7. The circuit configuration according to claim 6, wherein said
controllable resistors are field-effect transistors.
8. The circuit configuration according to claim 7, wherein said
controllable resistors formed by said field-effect transistors are
connected in parallel with one another.
9. The circuit configuration according to claim 7, wherein said
controllable resistors formed by said field-effect transistors are
connected in series with one another.
10. The circuit configuration according to claim 5, wherein each of said
integrators has a current flow and transistors that can be controlled
depending on the control signals, and said transistors control the current
flow for setting a transfer characteristic and a current drawn.
11. The circuit configuration according to claim 5, wherein each of said
integrators includes:
a positive supply voltage terminal;
a negative supply voltage terminal;
a first input terminal for receiving said third input signal;
a second input terminal for receiving said second input signal supplied by
a corresponding integrator of the other series circuit;
a first and second control terminal for receiving a first and second
control signal, respectively;
an output terminal for outputting an output signal;
a plurality of transistors having main current paths and including a first
transistor, a second transistor, a third transistor, a fourth transistor,
a fifth transistor, a sixth transistor, a sixth transistor, a seventh
transistor, an eighth transistor, a ninth transistor and a tenth
transistor;
a series circuit formed by said main current paths of said first and fourth
transistors connected between said positive supply voltage terminal and
said output terminal;
said first transistor having a control terminal connected to said first
input terminal;
said fourth transistor having a control terminal connected to said first
control terminal;
a series circuit formed by said main current paths of said second, third
and fifth transistors connected in parallel with said series circuit
formed by said first and fourth transistors;
said second transistor having a control terminal connected to said second
input terminal;
said third transistor having a control terminal connected to said output
terminal;
said fifth transistor having a control terminal connected to said first
control terminal; and
said sixth, seventh, eighth, ninth and tenth transistors connected to said
negative supply voltage terminal, said second control terminal, said first
input terminal, said second input terminal and said output terminal in a
mirror-symmetrical manner with respect to said first, second, third,
fourth and fifth transistors.
12. The circuit configuration according to claim 11, wherein said first,
second, third, fourth and fifth transistors are p-channel field-effect
transistors and said sixth, seventh, eighth, ninth and tenth transistors
are n-channel field-effect transistors.
13. The circuit configuration according to claim 11, wherein each of said
integrators further includes a third and fourth control terminal for
receiving a third and fourth control signal, respectively, as well as
eleventh and twelfth transistors, said eleventh transistor having a
control terminal and a main current path connected between said series
circuit formed by said first and fourth transistors and said output
terminal, and said control terminal of said eleventh transistor connected
to said third control terminal, and said twelfth transistor having a
control terminal and a main current path connected between said series
circuit formed by said sixth and tenth transistors and said output
terminal, and said control terminal of said twelfth transistor connected
to said fourth control terminal.
14. The circuit configuration according to claim 11, wherein said first
transistor and said sixth transistor are formed in each case by two
series-connected transistors having common control terminals.
15. The circuit configuration according to claim 5, including a control
circuit for generating said plurality of control signals depending on a
central control signal fed in externally and defining a switching
frequency of said integrators.
16. The circuit configuration according to claim 15, wherein said control
circuit for generating said plurality of control signals includes:
a positive supply voltage terminal;
a negative supply voltage terminal;
an input terminal for receiving the central control signal;
four output terminals including a first output terminal, a second output
terminal, a third output terminal and a fourth output terminal for
outputting a first, second, third and fourth control signal, respectively,
to said integrators;
two current sources;
a plurality of transistors having main current paths and including a first
transistor, a second transistor, a third transistor, a fourth transistor,
a fifth transistor, a sixth transistor, a seventh transistor, an eighth
transistor, a ninth transistor, a tenth transistor, an eleventh
transistor, a twelfth transistor, a thirteenth transistor and a fourteenth
transistor;
a series circuit formed by one of said two current sources and said main
current paths of said first and second transistors and connected between
said positive supply voltage terminal and said negative supply voltage
terminal;
a series circuit formed by said main current paths of said third and fourth
transistcrs and a second one of said two current sources and connected
between said positive supply voltage terminal and said negative supply
voltage terminal;
a series circuit formed by said main current paths of said fifth and sixth
transistors and connected between said positive supply voltage terminal
and said negative supply voltage terminal;
a series circuit formed by said main current paths of said seventh, eighth,
ninth, tenth, eleventh, twelfth, thirteenth and fourteenth transistors
connected between said positive supply voltage terminal and said negative
supply voltage terminal;
said twelfth transistor having a control terminal connected to said input
terminal and said second output terminal;
said ninth transistor having a control terminal connected to said first
output terminal;
said eleventh transistor having a control terminal connected to said third
output terminal;
said third output terminal connected between said one of said two current
sources and said series circuit of said first and second transistors; and
said tenth transistor having a control terminal connected to said fourth
output terminal, said fourth output terminal connected between said second
one of said two current sources and said series circuit of said third and
fourth transistors.
17. The circuit configuration according to claim 16, wherein said first,
second, fifth, sixth, eleventh, twelfth, thirteenth and fourteenth
transistors are n-channel field-effect transistors, and said third,
fourth, seventh, eighth, ninth and tenth transistors are p-channel
field-effect transistors.
18. The circuit configuration according to claim 16, including controllable
drivers having adjustable edge steepnesses and said output signals of said
integrators are passed via said controllable drivers.
19. The circuit configuration according to claim 18, wherein each of said
controllable drivers includes:
an input terminal for receiving an output signal of a corresponding
integrator;
an output terminal for outputting an output signal;
a first control terminal and a second control terminal for receiving the
first and second control signal, respectively, of said control circuit;
a positive supply voltage terminal;
a negative supply voltage terminal;
a plurality of transistors having main current paths and including a first
transistor, a second transistor, a third transistor, a fourth transistor,
a fifth transistor, a sixth transistor, a seventh transistor, an eighth
transistor, a ninth transistor, a tenth transistor, an eleventh
transistor, and a twelfth transistor;
a series circuit formed by said main current paths of said first, second
and third transistors connected to said positive supply voltage terminal;
a series circuit formed by said main current paths of said fifth and sixth
transistors connected between said positive supply voltage terminal and
said output terminal;
said fourth transistor connected by its main current path to a node between
said second and third transistors;
said first transistor having a control terminal connected to said input
terminal;
said second, third and sixth transistors each having a control terminal
connected to said first control terminal;
said fourth and fifth transistors each having a control terminal connected
to one another; and
said seventh, eighth, ninth, tenth, eleventh and twelfth transistors
connected up to said input terminal, said output terminal, said second
control terminal and said negative supply voltage terminal
mirror-symmetrically with respect to said first, second, third, fourth,
fifth and sixth transistors such that said third transistor is coupled to
said tenth transistor, and said ninth transistor is coupled to said fourth
transistor.
20. The circuit configuration according to claim 19, wherein said first,
second, third, fourth, fifth and sixth transistors are p-channel
field-effect transistors and said seventh, eighth, ninth, tenth, eleventh
and twelfth transistors are n-channel field-effect transistors.
21. A voltage regulator circuit, comprising:
an input terminal for receiving an input voltage;
an output terminal for providing a regulated output voltage;
a transistor having a control terminal and a main current path connected
between said input terminal and said output terminal;
a regulating amplifier having an input terminal receiving the regulated
output voltage and an output terminal outputting a control signal;
a capacitor connected between said control terminal of said transistor and
said output terminal of said regulating amplifier;
a controllable current source for discharging said capacitor; and
a charge pump for applying a voltage that can be controlled by said
regulating amplifier to said control terminal of said transistor, said
charge pump having a switching frequency that can be altered depending on
the control signal output by said regulating amplifier.
22. The voltage regulator circuit according to claim 21,
wherein said charge pump includes:
a first series circuit of n integrators having a first integrator and a
last integrator generating an output signal;
a first feedback path and a second feedback path;
one of said integrators of said first series circuit generating an output
signal and receiving, as an input signal, a summation signal formed from
said output signal of said one of said integrators of said first series
circuit fed back via said first feedback path, a second input signal and a
third input signal;
a second series circuit of n further integrators having a first integrator
and a last integrator generating an output signal;
a further first feedback path and a further second feedback path;
one of said integrators of said second series circuit generating an output
signal and receiving, as an input signal, a summation signal formed from
said output signal of said one of said integrators of said second series
circuit fed back via said further first feedback path, a second input
signal and a third input signal;
said output signal of said one of said integrators of said second series
circuit fed back via said second feedback path forming said second input
signal of said one of said integrators of said first series circuit;
said output signal of said one of said integrators of said first series
circuit fed back via said further second feedback path forming said second
input signal of said one of said integrators of said second series
circuit;
said output signal of said last integrator of said second series circuit
forming said third input signal of said one of said integrators
corresponding to said first integrator of said first series circuit;
said output signal of said last integrator of said first series circuit
supplying said third input signal of said one of said integrators
corresponding to said first integrator of said second series circuit;
said third input signal for remaining integrators of said first series
circuit supplied by an output signal of a preceding integrator of said
integrators of said first series circuit;
said third input signal for remaining integrators of said second series
circuit supplied by an output signal of a preceding integrator of said
integrators of said second series circuit; and
said integrators of said first series circuit and said second series
circuit having outputs that can be picked off outputting output signals
phase-shifted with respect to one another.
23. The voltage regulator circuit according to claim 21,
wherein said transistor is a DMOS field-effect transistor.
24. The voltage regulator circuit according to claim 21,
wherein said regulating amplifier is a Class B amplifier with a low
quiescent current.
25. The voltage regulator circuit according to claim 24, including a
further current source coupled to said control terminal of said
transistor, in that, given an input voltage of a specific magnitude, said
regulating amplifier generates the control signal for said charge pump,
said control signal corresponding to a switching frequency zero, and said
further current source configured and disposed such that it maintains a
sufficiently high control charge for said control terminal of said
transistor even in a case of the control signal of said regulating
amplifier corresponding to the switching frequency zero.
Description
BACKGROUND OF THE INVENTION
Field of the Invention
The present invention relates to a circuit configuration for a charge pump
and also to a voltage regulator in which a circuit configuration of this
type is used in a charge pump.
The operation of various electronic circuits necessitates the use of
voltage regulators that ensure that the operating voltage is as far as
possible free from interference and stabilized. Thus, by way of example,
electronic units in motor vehicles require voltage regulators which
produce an interference-free, stabilized output voltage (of the order of
magnitude of approximately 5 V) from the operation voltage (between 12 and
15 V) on which interference is superposed. The voltage regulators used in
this field of application must have a very low current consumption (<100
.mu.A) since the majority of electronic motor vehicle systems remain
connected to the on-board electrical system of the motor vehicle even when
the motor vehicle is turned off, but the vehicle battery must not be
discharged in his case. Furthermore, the voltage regulators used should
enable the smallest possible voltage drop between the input and output
voltages, in order that an output voltage which is still sufficiently high
can be maintained even in the event of dips in the supply voltage or
on-board electrical system voltage to 5.5 V, for example.
Charge pumps are frequently used for driving power MOS transistors. To
ensure that the charge pump circuit used in this context manages with the
smallest possible integrated capacitors, it must, however, be operated at
the highest possible frequency, as a result of which, however, the problem
of electromagnetic radiation arises. In order to counteract this problem,
Published, European Patent Application EP 0 633 662 A1, for example, has
proposed a charge pump including a circuit configuration which generates a
variety of signals which are nested in one another, in each case two of
these signals having flattened edges having precisely complementary
profiles with respect to one another, which edges lead to a lower degree
of electromagnetic radiation of the charge pump. The circuit configuration
described in Published, European Patent Application EP 0 633 662 A1
essentially corresponds to a ring oscillator with two series circuits
formed by a plurality of driver stages, the output of the last driver
stage of one series circuit being connected to the input of the first
driver stage of the other series circuit. The outputs of the driver stages
of the individual series circuits are connected to one another via holding
stages formed, by way of example, by in each case two inverters connected
up oppositely to one another. Inverted output signals can be picked off at
the outputs of the individual driver stages. As has already been
explained, this circuit configuration can be used to suppress emitted
interference, which is of considerable importance in the case of voltage
regulators. However, the charge pump described in Published, European
Patent Application EP 0 633 662 A1 generates square-wave output signals,
as a result of which interference continues to be emitted in principle,
albeit in a weakened form.
SUMMARY OF THE INVENTION
It is accordingly an object of the invention to provide a circuit
configuration for a charge pump and voltage regulator circuit having a
circuit configuration of this type that overcomes the above-mentioned
disadvantages of the prior art devices of this general type, in which an
improved circuit configuration for the charge pump which, in particular,
enables a reduced current consumption and is suitable for use in voltage
regulators on smart power ICs in accordance with DMOS technology.
Furthermore, the intention is for the circuit configuration to have
improved electromagnetic compatibility (EMC). In addition, the present
invention is based on the object of providing an improved voltage
regulator that has a reduced current consumption for a predetermined
maximum output voltage.
With the foregoing and other objects in view there is provided, in
accordance with the invention, a circuit configuration for a charge pump,
including:
a first series circuit of n integrators having a first integrator and a
last integrator generating an output signal;
a first feedback path and a second feedback path;
one of the integrators of the first series circuit generating an output
signal and receiving, as an input signal, a summation signal formed from
the output signal of the one of the integrators of the first series
circuit fed back via the first feedback path, a second input signal and a
third input signal;
a second series circuit of n further integrators having a first integrator
and a last integrator generating an output signal;
a further first feedback path and a further second feedback path;
one of the integrators of the second series circuit generating an output
signal and receiving, as an input signal, a summation signal formed from
the output signal of the one of the integrators of the second series
circuit fed back via the further first feedback path, a second input
signal and a third input signal;
the output signal of the one of the integrators of the second series
circuit fed back via the second feedback path forming the second input
signal of the one of the integrators of the first series circuit;
the output signal of the one of the integrators of the first series circuit
fed back via the further second feedback path forming the second input
signal of the one of the integrators of the second series circuit;
the output signal of the last integrator of the second series circuit
forming the third input signal of the one of the integrators corresponding
to the first integrator of the first series circuit;
the output signal of the last integrator of the first series circuit
supplying the third input signal of the one of the integrators
corresponding to the first integrator of the second series circuit;
the third input signal for remaining integrators of the first series
circuit supplied by an output signal of a preceding integrator of the
integrators of the first series circuit;
the third input signal for remaining integrators of the second series
circuit supplied by an output signal of a preceding integrator Df the
integrators of the second series circuit; and
the integrators of the first series circuit and the second series circuit
having outputs that can be picked off outputting output signals
phase-shifted with respect to one another.
According to the present invention, the circuit configuration has two
series circuits formed by a plurality of integrators which are each
coupled to one another via corresponding feedback paths, with the result
that corresponding sinusoidal output signals which are offset in phase
with respect to one another can be generated at the outputs of the
individual integrators. This being advantageous with regard to the
electromagnetic compatibility (EMC), in contrast to the phase-shifted
square-wave pulses known from Published, European Patent Application EP 0
633 662 A1. The circuit configuration advantageously contains four
integrators that, consequently, generate four approximately sinusoidal
output signals which are offset in phase in each case by 90.degree.
relative to one another.
The integrators are advantageously realized in the form of controlled
inverters with a plurality of inputs, with the result that specific
properties of the integrators can be altered by the application of
corresponding control signals. In this case, this aspect of the invention
is based, in particular, on the insight that the current consumption of
the circuit configuration or of the entire voltage regulator can be
distinctly reduced if the (circuit) frequency of the charge pump is
reduced to an extent such that the current output of the charge pump
decreases to a required value. Consequently, by driving the controllable
integrators of the circuit configuration, it is possible to generate for
example four approximately sinusoidal output signals which are offset in
phase in each case by 90.degree. relative to one another and whose
frequency can be altered with the aid of external control signals.
The use of the circuit configuration according to the invention in a
voltage regulator of the type described in the introduction is
particularly advantageous if the regulating amplifier used in the process
is configured as a so-called
Class B amplifier with a low quiescent current and, given a sufficiently
high input voltage, generates a control signal for the charge pump of the
voltage regulator which corresponds to the output frequency zero. In this
context, use is made of an additional current source that maintains the
gate charge necessary for the DMOS actuating transistor in this case.
The variable-frequency charge pump proposed in connection with the voltage
regulator according to the invention can be used, in particular, for
so-called high side switches, in which rapid switching on and, once the
switch has been switched on, a low degree of interference emission
together with a low current consumption are required.
Other features which are considered as characteristic for the invention are
set forth in the appended claims.
Although the invention is illustrated and described herein as embodied in a
circuit configuration for a charge pump and voltage regulator circuit
having a circuit configuration of this type, it is nevertheless not
intended to be limited to the details shown, since various modifications
and structural changes may be made therein without departing from the
spirit of the invention and within the scope and range of equivalents of
the claims.
The construction and method of operation of the invention, however,
together with additional objects and advantages thereof will be best
understood from the following description of specific embodiments when
read in connection with the accompanying drawings.
BRIEF DESCRIPTION OF THE DRAWINGS
FIG. 1 is a diagrammatic, block diagram of a preferred exemplary embodiment
of a voltage regulator according to the invention;
FIG. 2a is a block diagram of a doubling charge pump;
FIGS. 2b-2c are graphs elucidating why a current consumption can be reduced
by reducing a frequency of the charge pump;
FIG. 3 is a circuit diagram of a preferred exemplary embodiment of a
circuit configuration for the charge pump;
FIG. 4 is a circuit diagram of an integrator illustrated in FIG. 3;
FIG. 5 is a circuit diagram of a driver shown in FIG. 3;
FIG. 6 is a circuit diagram of a circuit for generating control signals for
the drivers and the integrators shown in FIGS. 3-5;
FIG. 7 is a circuit diagram of the voltage regulator according to a further
preferred exemplary embodiment; and
FIGS. 8-10 are circuit diagrams of prior art voltage regulators.
DESCRIPTION OF THE PREFERRED EMBODIMENTS
In all the figures of the drawing, sub-features and integral parts that
correspond to one another bear the same reference symbol in each case.
Referring now to the figures of the drawing in detail and first,
particularly, to FIG. 8 thereof, there is shown a simplified block diagram
of a known voltage regulator circuit with a pnp actuating transistor T
being used as the actuator. With the aid of the actuating transistor T, an
input voltage U.sub.e is converted into an output voltage U.sub.a which is
as far as possible free from interference and stabilized. One input
terminal of a regulating amplifier RV monitors the output voltage U.sub.a
and generates an actuating signal for a controllable current source I on
the basis of a comparison with a reference voltage U.sub.ref applied to
the other input terminal, which controllable current source generates a
control current for the actuating transistor T. As a result of the
interaction of the actuating transistor T with the regulating amplifier RV
and the controllable current source I, the desired stabilized output
voltage U.sub.a can be produced. The use of the pnp actuating transistor T
shown in FIG. 8 is advantageous on account of the cost-effective bipolar
technology used in this context, which enables powerful pnp transistors.
However, the trend in future electronic units is toward "Systems on
Silicon", that is to say the intention is for a chip to be able to include
as many different functions as possible in order that a unit can be
constructed from as few chips as possible. Endeavors are being made,
therefore, to integrate voltage regulators on so-called "Smart Power
Chips" which are fabricated in accordance with a hybrid technology, which
are optimized, on the one hand, for DMOS power transistors and CMOS logic,
and, in addition, have bipolar components. A further aim is to keep the
external circuitry of the voltage regulators as small as possible.
The voltage regulator shown in FIG. 8 and containing the pnp bipolar
transistor T always requires a dedicated chip, however. In addition, the
low transition frequency of integrated lateral pnp transistors means that
the relatively large output capacitor C, Likewise shown in FIG. 8, is
necessary, which output capacitor has a low series resistance over the
widest possible temperature range. By way of example, the capacitor C may
be formed by a tantalum electrolytic capacitor having a capacitance of
between 10 and 22 .mu.F.
For smart power technologies, therefore, use has been made heretofore of
voltage regulators having a DMOS actuating transistor with an associated
charge pump for producing the requisite high gate voltage. FIG. 9 shows a
block diagram of a known DMOS voltage regulator circuit. It is evident
from FIG. 9 that an n-channel DMOS field-effect transistor is used as the
actuating transistor T, the gate terminal of which field-effect transistor
is driven by controllable current sources I1 and I2. The two current
sources I1 and I2 are driven, for their part, by the output signal of the
regulating amplifier RV, the current source I1 being set by the output
signal of the regulating amplifier RV in an inverted manner with respect
to the current source I2. Furthermore, FIG. 9 illustrates a charge pump
LP, which ensures that an increased gate voltage is applied to the gate
terminal of the actuating transistor T.
The voltage regulator shown in FIG. 9 has the disadvantage, however, in
that it has a very high current consumption (for example a few mA) on
account of the quiescent current consumption of the charge pump LP, with
the result that in motor vehicle applications, for example, the voltage
regulator must also be switched off when the vehicle is turned off.
However, the use of n-channel DMOS transistors is advantageous in voltage
regulators because, in general, pnp or p-channel MOS transistors are not
suitable for smart power chips since they can only be operated with low
current densities given application of the smart power technology.
FIG. 10 shows a block diagram of a further known voltage regulator circuit,
as is disclosed in U.S. Pat. No. 5,654,628, for example. This voltage
regulator includes an input terminal for receiving an (unregulated) input
voltage U.sub.e and an output terminal for providing a (regulated) output
voltage U.sub.a, an n-channel DIOS field-effect transistor T being
connected as the actuator between the input and output terminals.
Furthermore, use is again made of the regulating amplifier RV, which
receives the regulated output voltage U.sub.a at one of its inputs and
generates a corresponding output signal for the controllable current
sources I1 and I2. A capacitor C1 is connected between the output terminal
of the regulating amplifier RV and the control terminal of the transistor
T. The current source I2 corresponds to the internal current source of the
charge pump LP, which, owing to corresponding driving by the regulating
amplifier RV, generates an increased output voltage for the transistor T,
in order to boost the gate voltage of the transistor T. The current source
I1 serves for discharging the capacitor C1, in which case, in the event of
a drop in the output voltage U.sub.a, an increased output signal of the
regulating amplifier RV ensures that the current of the current source I1
is reduced owing to the inverted driving by the regulating amplifier RV.
Furthermore, the output signal of the regulating amplifier RV is fed to
the transistor T via the capacitor C1, with the result that the signal
U.sub.a at the output terminal is increased owing to the relatively
constant voltage drop along the gate-source path of the transistor T. With
the aid of the capacitor C1, rapid output-side load changes of the voltage
regulator can be compensated for, and the voltage regulator can also
operate stably with a small output capacitance C2 (for example with a 100
.mu.F ceramic capacitor).
A voltage regulator according to the invention that essentially corresponds
to the voltage regulator shown in FIG. 10 is shown in FIG. 1. The voltage
regulator shown in FIG. 1 includes the input terminal for receiving the
unregulated input voltage U.sub.e and also the output terminal for
providing the (regulated) output voltage U.sub.a. The actuating transistor
T is connected by its main current path or its load path between the input
and output terminals. In accordance with FIG. 1, the transistor T is
configured, in particular, in the form of an N-channel DMOS field-effect
transistor. At its inverted input, the regulating amplifier RV monitors
the output voltage U.sub.a provided at the output terminal and compares
the output voltage with the reference voltage U.sub.ref applied to its
further input. The capacitor C1 is connected between the output of the
regulating amplifier RV and the control or gate terminal of the actuating
transistor T. The current source I serves for discharging the capacitor C1
and is controlled in an inverted manner by the output signal of the
regulating amplifier RV. Furthermore, provision is made of the charge pump
LP that generates an increased voltage for the control terminal of the
actuating transistor T. An output capacitor C2 is provided at the output
of the voltage regulator, and the output voltage U.sub.a is output via the
capacitor C2.
In contrast to the voltage regulator shown in FIG. 10, however, in
accordance with the exemplary embodiment shown in FIG. 1, the frequency of
the charge pump LP is set by the output signal of the regulating amplifier
RV. That is to say the charge pump LP is a voltage-controlled,
variable-frequency charge pump (VCO charge pump), the frequency of the
charge pump LP being adapted to the respective requirements for the
purpose of reducing the current consumption.
This will be explained in more detail below with reference to the diagrams
illustrated in FIGS. 2a-2c.
FIG. 2a shows a greatly simplified block diagram of a doubling charge pump
which, in accordance with FIG. 1 receives a voltage U.sub.LPe as an input
voltage and outputs an output voltage U.sub.LPa. In accordance with FIG.
2a the charge pump can be represented as a combination of a ring
oscillator OSC with a downstream driver TR. The output signals of the
driver TR being output via a diode-capacitor network having diodes D1 and
D2 as well as a pump capacitor C. Furthermore, FIG. 2a illustrates a
capacitor Cs, which corresponds to the unavoidable parasitic substrate
capacitance of the pump capacitors used in the charge pump. The parasitic
substrate capacitance being governed by the monolithic integration. The
following relationships exist between the current I.sub.LPe taken up by
the doubling charge pump and the current I.sub.LPa that is output:
I.sub.LPe =C.sub.s .multidot.f.multidot.U.sub.LPe
+2.multidot.C.multidot.f.multidot.(2.multidot.U.sub.LPe -U.sub.LPa
-2.multidot.U.sub.D)
I.sub.LPa =C.multidot.f.multidot.(2.multidot.U.sub.LPe -U.sub.LPa
-2.multidot.U.sub.D)
FIG. 2b illustrates the profile of the current I.sub.LPe that is taken up
and the current I.sub.LPa that is output, as a function of the output
voltage U.sub.LPa of the doubling charge pump. It is evident from the
above formulas that the charging current necessary for periodic charging
and discharging of the parasitic capacitances constitutes a main
proportion of the no-load current consumption of a charge pump. Further
proportions of the quiescent current consumption of the charge pump are
produced, moreover, by shunt currents of the driver stages TR and by the
current consumption of the oscillator OSC.
In a supplementary manner, FIG. 2c illustrates the efficiency as a function
of the output voltage U.sub.LPa. In this case, it is evident from FIG. 2c
that, in the event of no-load operation of the charge pump, the efficiency
n is equal to zero and has a maximum at an output voltage ULPa=7 V given
an input voltage ULpe=5 V, a useful. capacitance C=5 pF and a substrate
capacitance Cs=1.2 pF. If only a low power output is required from the
charge pump, this being the case for example with a settled regulating
circuit and with low loading on the voltage regulator, the charge pump
should not operate under no load for this reason, which would initially be
obvious, or the input voltage U.sub.LPe of the charge pump should be
reduced, which is proposed in accordance with U.S. Pat. No. 5,654,628 (see
FIG. 10).
Rather, it is advantageous to reduce, instead, the frequency of the charge
pump to an extent such that the current output of the charge pump
decreases to the desired or required value, in order to obtain a very low
current consumption. Therefore, according to FIG. 1, the charge pump LP is
configured in such a way that its frequency is variable depending on the
output signal of the regulating amplifier RV. In this case, the charge
pump LP may be configured as a multiphase charge pump which generates a
plurality of sinusoidal output signals which are offset in phase relative
to one another. It being possible to alter the frequency of these output
signals with the aid of the control signal VCO of the regulating amplifier
RV. The highest frequency that can be generated by the charge pump LP
should advantageously be just as high as the frequency that can be
generated with the aid of an oscillator based on inverter transit times
(see Published European Patent Application EP 0 633 662 A1) described in
the introduction.
The use of variable-frequency charge pumps has already been disclosed, in
principle, in IEEE Journal of Solid State Circuits, Vol. 32, No. 6, June
1997, "Efficiency Improvement in Charge Pump Circuits". In particular, the
document describes the problem whereby charge pumps use transistor
circuits which are operated at a specific switching frequency. The
switching frequency being defined, according to the known prior art, in
such a way that a sufficiently high output signal level can be obtained
even with a maximum output load of the charge pump. The consequence of
this being, however, that in the event of low loading on the charge pump,
a considerable supply current is unnecessarily required, the consequence
being a considerable energy consumption particularly in the case of
systems which, for the most part, are operated without a load.
Accordingly, this document proposed varying the switching frequency of the
charge pump depending on the load, with the result that switching
frequency of the charge pump is reduced in the event of a smaller load.
This results in an improved performance, a lower current consumption and
less stringent breakdown voltage requirements for the charge pump. This
document does not, however, describe the use of the charge pump in voltage
regulator circuits, with the result that this document also does not make
any proposal as to how the driving of the charge pump could be effected in
a voltage regulator circuit, in order to vary the frequency of the charge
pump.
By contrast, FIG. 3 shows a circuit diagram of a preferred exemplary
embodiment of the variable-frequency charge pump LP which is illustrated
in FIG. 1 and which additionally fulfills the previously known
preconditions. The circuit shown in FIG. 3 includes two series circuits
having a plurality of jointly driven integrators IN1-IN4, the individual
integrators IN1-IN4 being connected up to one another via feedback paths.
In particular, FIG. 3 illustrates an example of a doubling multiphase
charge pump, each series circuit having two controlled integrators IN1,
IN2 and IN3, IN4, respectively. The output of the last integrator of each
series circuit is passed to the input of the first integrator of the other
series circuit. Furthermore, the output of each integrator is passed to
its input via a first feedback path having the gain factor K1 and to the
input of the corresponding integrator of the other series circuit via a
second feedback path having the gain factor K2. The input signal of each
integrator IN1-IN4 corresponds to the output signal of an adder ADD1-ADD4.
Consequently, the input signal of the integrators IN1 and IN3 corresponds
to the summation signal including the output signal of the integrator IN4
and of the integrator IN2, respectively, the output signal of the
integrator IN1 and IN3, respectively, multiplied by the gain factor K1 and
the output signal of the integrator IN3 and IN1, respectively, multiplied
by the gain factor K2. On the other hand, the input signal of the
integrators IN2 and IN4 corresponds to the summation signal including the
output signal of the integrator IN1 and IN3, respectively, the output
signal of the integrator IN2 and IN4, respectively, multiplied by the gain
factor K1 and the output signal of the integrator IN4 and IN2,
respectively, multiplied by the gain factor K2.
The gain factors of the feedback paths are set in such a way that the
following holds true: K2>K1, K2-K1<1 (oscillation condition) and K1+K2>1
(stability condition).
According to FIG. 3, the individual integrators IN1-IN4 are driven
depending on the control signal VCO of the regulating amplifier RV shown
in FIG. 1. Approximately sinusoidal output signals which are offset in
phase in each case by 90.degree. with respect to one another are picked
off at the outputs of the individual integrators INI-IN4 and are passed
via driver stages TR1-TR4. It being the case that these driver stages
TR1-TR4 are also preferably set depending on the control signal VCO, in
order, for example, to alter the edge steepness of the drivers TR1-TR4.
The output signals are combined, by a diode-capacitor network having
capacitors C3-C6 and diodes D1-D8, with the input voltage U.sub.Lpe
applied to the charge pump, in order finally to obtain the increased
output voltage U.sub.LPa.
Although the circuit configuration is configured, according to FIG. 3, with
controllable Lntegrators IN1-IN4 and controllable drivers TR1-TR4, the
circuit configuration shown in FIG. 3 can, of course, also be configured
without a control capability for the integrators and/or drivers, since in
this case too, sinusoidal signals which are offset in phase with respect
to one another can be picked off at the outputs of the integrators, this
being distinctly advantageous, with regard to the electromagnetic
compatibility (EMC) in comparison with the phase-shifted square-wave
pulses disclosed in Published, European Patent Application EP 0 633 662
A1.
The controlled integrators IN1-IN4 shown in FIG. 3 can be realized as
controlled inverters with a plurality of inputs, in particular a plurality
of control inputs. In this case, the controllable transfer characteristic
of the inverter forms, together with the wiring capacitance, in each case
a controllable integrator. FIG. 4 shows a possible circuit for a
controllable integrator.
According to FIG. 4, the circuit contains a positive supply voltage
terminal VDD, a negative supply voltage terminal VSS, a first input
terminal IN1 for the first input signal of the corresponding adder, a
second input terminal IN2 for the second input signal of the corresponding
adder and also four control terminals VP, VPP, VNN and VN, to which
corresponding control signals are to be applied. The integrator or
inverter shown in FIG. 4 is constructed using a combination circuit of
p-channel and n-channel junction field-effect transistors. In this case,
the transistors MN1-MN3 and MP1-MP3 form controlled resistors which are
dependent on the input signals. The resistors can optionally be connected
in parallel or in series, connection in parallel enabling the maximum
possible frequency of the oscillator of the charge pump to be increased
and connection in series enabling the current consumption of the
oscillator to be reduced. In accordance with the exemplary embodiment
shown in FIG. 4, a combination of connection in parallel and connection in
series was chosen, which turned out to be a favorable compromise, in each
case two transistors MN2, MN3 and MP2, MP3, respectively, being connected
in series, while a further transistor MN1 and MP1, respectively, is
disposed in parallel therewith. For reasons of manufacturability, however,
according to FIG. 4 the transistors MN1 and MP1 which are not connected in
series are also likewise configured in the form of a series circuit
respectively including two transistors MN1A and MN1B, and MP1A and MP1B.
The requisite gain factors of the individual inputs can be set by
different channel lengths and channel widths of the transistors.
Furthermore, the circuit shown in FIG. 4 includes transistors MN4, MN5 and
MN5, MP5, which serve as current source transistors and, depending on the
control signals VP and VN, control the total current flow via the inverter
and hence the transfer characteristic of the inverter. In addition, it is
optionally possible to insert cascade transistors MP6 and MN6, which are
controlled by the control signals VPP and VNN, respectively, and serve to
extend the frequency range. What is evident overall from FIG. 4 is that
the inverter circuit consists of essentially mirror-symmetrical p-channel
and n-channel MOS transistor regions.
As has already been explained with reference to FIG. 3, the individual
drivers TR1-TR4 can also be set depending on the output signal of the
regulating amplifier RV, it being particularly expedient, for minimum
interference radiation, to set the edge steepnesses of these output
drivers proportionally to the oscillator frequency of the charge pump.
FIG. 5 shows an example of a shunt current-free driver circuit having the
features mentioned, it being possible for the edge steepness to be set, in
particular, depending on the control signals VN, VP (cf. FIG. 4). The
driver circuit shown in FIG. 5 also once again includes the combination of
p-channel and n-channel MOS junction transistors MP7-MP12 and MN7-MN12,
respectively, the region having the p-channel transistors being
constructed essentially mirror-symmetrically with respect to the region
having the n-channel transistors.
The integrator shown in FIG. 4 and also the driver shown in FIG. 5 are
driven depending on the control signals VP, VPP, VNN and VN which are
derived from the output signal VCO of the regulating amplifier RV (see
FIG. 1). Accordingly, the charge pump shown in FIG. 3 must have a control
circuit which generates the corresponding control signals VP, VPP, VNN and
VN, respectively, depending on the output signal VCO of the regulating
amplifier RV. A corresponding control circuit is illustrated by way of
example in FIG. 6, which once again has a combination of p-channel
junction field-effect transistors MP13-MP1 and n-channel junction
field-effect transistors MN13-MN20. Furthermore, the control circuit
includes internal current sources I3 and I4. The more detailed circuitry
connections can be inferred from FIG. 6. Since, according to FIG. 3, the
individual integrators IN1-IN4 and drivers TR1-TR4 are intended to be set
jointly depending on the control signal VCO of the regulating amplifier
RV, it suffices to provide a central control circuit, corresponding to the
circuit shown in FIG. 6, in the charge pump, which generates the control
signals VPP, VP, VN and VNN depending on the control signal VCO. As an
alternative, however, it is also conceivable for the circuit shown in FIG.
6 to be integrated in each of the circuits shown in FIGS. 4 and 5, that is
to say in each integrator and driver circuit.
Finally, FIG. 7 shows a further preferred exemplary embodiment of the
voltage regulator according to the invention, in which case it is possible
to use, in particular, the variable-frequency charge pump LP (VCO charge
pump) which is shown in FIG. 3 and whose frequency is once again set
depending on the output signal VCO of the regulating amplifier RV, in
order, consequently, to adapt the frequency of the charge pump LP to the
instantaneous load or to the required current output and, consequently, to
minimize the current consumption of the charge pump. Furthermore,
according to FIG. 7, the regulating amplifier RV is configured as a
so-called Class B amplifier with a low quiescent current and generates,
given a sufficiently high input voltage (for example greater than 8 V),
such a control signal VCO for the charge pump LP which corresponds to the
frequency 0. In order to maintain the gate charge required for the
actuating transistor T in this case, too, according to FIG. 7 a further
current source I2 is provided, which is coupled to the gate terminal of
the transistor T via a diode D. The current consumption of the voltage
regulator can be reduced further with the aid of the voltage regulator
circuit shown in FIG. 7.
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