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United States Patent |
6,034,516
|
Goerke
,   et al.
|
March 7, 2000
|
Soft-start switch with voltage regulation and current limiting
Abstract
A MOSFET, an op-amp, a comparator circuit, and voltage dividers with
capacitors are employed in combination to effectuate a soft-start switch
with current limiting. The transconductance of the MOSFET is employed so
that no sense resistor is required. The MOSFET and op-amp are configured
as a closed-loop feedback circuit in which the output of the op-amp is
coupled to the gate of the MOSFET and the inverting input of the op-amp is
coupled to the output of the soft-start switch via a voltage divider. A
first RC circuit provides a voltage to the non-inverting input of the
op-amp which can be triggered to gradually rise from a value close to zero
to some reference voltage so as to soft-start a load. Current limiting
means are effectuated by a comparator circuit and voltage dividers with
capacitors. The current limiting means brings the MOSFET to an OFF state
and the non-inverting input of the op-amp close to zero volts if the
op-amp charges a second RC circuit so that the voltage drop across its
capacitor exceeds a pre-determined limit-reference, and also, once the
current limiting means brings the MOSFET to the OFF state, the current
limiting means allows the soft-start switch to begin a soft-start power-up
after a pre-determined time dependent upon the time constant of the second
RC circuit.
Inventors:
|
Goerke; Uli B. (Boylston, MA);
Pieper; Mark (Marlboro, MA)
|
Assignee:
|
Data General Corporation (Westborough, MA)
|
Appl. No.:
|
893803 |
Filed:
|
July 11, 1997 |
Intern'l Class: |
G05F 001/40 |
Field of Search: |
323/238,282,285,901
363/49,21
322/25
|
References Cited
U.S. Patent Documents
4161760 | Jul., 1979 | Valentine | 361/18.
|
4924170 | May., 1990 | Henze | 323/285.
|
5015921 | May., 1991 | Carlson et al.
| |
5045771 | Sep., 1991 | Kislovski.
| |
5063303 | Nov., 1991 | Sackman et al. | 307/296.
|
5257156 | Oct., 1993 | Kirkpatrick.
| |
5376831 | Dec., 1994 | Chen.
| |
5528132 | Jun., 1996 | Doluca.
| |
5619127 | Apr., 1997 | Warizaya.
| |
Other References
"Programmable Electronic Circuit Breaker" Unitrode, UCC 3912, pp. 6-437
6-441.
|
Primary Examiner: Hecker; Stuart N.
Attorney, Agent or Firm: Bromberg & Sunstein LLP
Parent Case Text
This is a continuation of application Ser. No. 08/690,540 filed on Jul. 31,
1996, now U.S. Pat. No. 5,698,973.
Claims
We claim:
1. A voltage regulator to limit a pass current from a power source to a
load and to regulate a load voltage applied to the load, the voltage
regulator comprising:
a voltage-controlled current device having a first terminal, a second
terminal coupled to the power source, and a third terminal coupled to the
load, wherein the pass current flows between the second and third
terminals and there is a transconductance relationship between the pass
current and the voltage difference between the first and third terminals;
a control circuit responsive to the load voltage, and having an input and
having an output coupled to the first terminal of the voltage-controlled
current device so as to regulate the load voltage in accordance with a
voltage at the input to the control circuit; and
a current limit circuit, coupled to the input of the control circuit and
coupled to the control circuit output and the third terminal of the
voltage-controlled current device so as to limit the pass current
responsive to a voltage difference between the voltage of the control
circuit output and the voltage of the third terminal of the
voltage-controlled current device.
2. The voltage regulator as set forth in claim 1, wherein the control
circuit comprises:
an op-amp with its output coupled to the output of the control circuit and
its non-inverting input coupled to the input of the control circuit;
a resistor connecting the first terminal of the voltage-controlled current
device to the output of the op-amp; and
a voltage divider circuit coupling the second terminal of the
voltage-controlled current device to the load and coupled to the inverting
input of the op-amp to provide negative feedback.
3. The voltage regulator of claim 1, further comprising:
a first voltage divider circuit, coupled to the voltage-controlled current
device, and having a first node with a first voltage; and
a second voltage divider circuit, coupled to the output of the control
circuit and the voltage-controlled current device, and having a second
node with a second voltage; wherein the current limit circuit is
responsive to the first and second voltages of the first and second nodes
so as to drive the voltage-controlled current device into an OFF state
when the second voltage exceeds the first voltage.
4. The voltage regulator of claim 3, wherein the voltage-controlled current
device is a MOSFET.
5. The voltage regulator as set forth in claim 3, wherein the first voltage
divider circuit includes:
a first resistor connecting the third terminal of the voltage-controlled
current device to the first node; and
a second resistor connecting the first node to a voltage source; and the
second voltage divider circuit includes:
a third resistor connecting a third node to the second node; and
a fourth resistor connecting the second node to ground.
6. The voltage regulator of claim 5, further comprising:
a diode connecting the output of the control circuit with the third node;
and
a capacitor connecting the third terminal of the voltage-controlled current
device to the third node.
7. The voltage regulator as set forth in claim 3, wherein the current limit
circuit includes:
a comparator responsive to the first and second voltages;
a first diode coupling the output of the comparator to the input of the
control circuit to provide to the input of the control circuit a first
high impedance to ground when the first voltage is greater than the second
voltage and to provide a first low impedance to ground when the second
voltage is greater than the first voltage; and
a second diode connecting the output of the comparator to the first node to
provide positive feedback.
8. The voltage regulator as set forth in claim 7, wherein the current limit
circuit further includes a third diode coupling the output of the
comparator with the first terminal of the voltage-controlled current
device to provide to the first terminal a second low impedance to ground
when the second voltage is greater than the first voltage so as to force
the voltage-controlled current device into the OFF state, and provides to
the first terminal of the voltage-controlled current device a second high
impedance to ground when the first voltage is greater than the second
voltage.
9. The voltage regulator as set forth in claim 8, wherein the current limit
circuit further includes a capacitor connecting the first node to the
third terminal of the voltage-controlled current device.
10. A voltage regulator with current limiting for providing pass current to
a load, the voltage regulator having an input and an output, the voltage
regulator comprising:
a voltage-controlled current device to control the pass current, with a
first terminal, a second terminal, and a third terminal, wherein the
second terminal is coupled to the input of the voltage regulator and the
third terminal is coupled to the output of the voltage regulator, wherein
the pass current flows between the second and third terminals and is
responsive to the voltage difference between the first and third
terminals;
a control circuit to control the voltage at the output of the voltage
regulator, with an input and with an output coupled to the first terminal
of the voltage-controlled current device, wherein the coupling between the
control circuit and the output of the voltage regulator is such as to
provide negative feedback;
voltage means, coupled to the output of the control circuit and the third
terminal of the voltage-controlled current device, for providing a first
voltage at a first node and a second voltage at a second node, where the
first voltage is a first function of a source voltage and of the voltage
regulator output voltage and the second voltage is a second function of a
third voltage at a third node, where the third node is coupled to the
output of the control circuit and the third terminal of the
voltage-controlled current device; and
a current limit circuit to cause the control circuit to drive the
voltage-controlled current device into an OFF state, so as to limit the
pass current, when the first voltage at the first node is less than the
second voltage at the second node, where the first and second nodes are
coupled to the current limit circuit.
11. The voltage regulator as set forth in claim 10, wherein the first and
second functions are non-decreasing.
12. The voltage regulator as set forth in claim 10, wherein the voltage
means includes:
a first resistor connecting the third terminal of the voltage-controlled
current device to the first node;
a second resistor connecting the first node to a voltage source providing
the source voltage;
a third resistor connecting the third node to the second node; and
a fourth resistor connecting the second node to ground.
13. The voltage regulator as set forth in claims 12, further comprising:
a first diode connecting the third node to the output of the control
circuit; and
a first capacitor connecting the third node to the third terminal of the
voltage-controlled current device.
14. The voltage regulator as set forth in claim 13, wherein the current
limit circuit provides to the input of the control circuit either a first
low impedance to ground when the second voltage is greater than the first
voltage, or a first high impedance to ground when the first voltage is
greater than the second voltage, wherein providing the first low impedance
causes the control circuit to force the voltage-controlled current device
into the OFF state.
15. The voltage regulator as set forth in claim 14, wherein the current
limit circuit further comprises:
a comparator,
a second diode coupling the output of the comparator to the input of the
control circuit, to provide to the input of the control circuit the first
high impedance to ground when the first voltage is greater than the second
voltage, and to provide the first low impedance to ground when the second
voltage is greater than the first voltage;
a third diode coupling the output of the comparator to the first node to
provide positive feedback; and
a fourth diode coupling the output of the comparator to the first terminal
of the voltage-controlled current device to provide a second low impedance
to ground when the second voltage is greater than the first voltage so as
to drive the voltage-controlled current device into the OFF state, and to
provide a second high impedance to ground when the first voltage is
greater than the second voltage.
16. The voltage regulator as set forth in claim 15, wherein the control
circuit includes:
an op-amp with an inverting input responsive to the soft-start output
voltage so as to provide negative feedback, a non-inverting input
connected to the input of the control circuit, and an output; and
a fifth resistor connecting the output of the control circuit to the first
terminal of the voltage-controlled current device.
17. The voltage regulator as set forth in claim 16, wherein the
voltage-controlled current device is a MOSFET.
18. A method for limiting pass current supplied to a load by a power
source, the method comprising the steps of:
providing a voltage-controlled current device having a first terminal, a
second terminal coupled to the power source, and a third terminal coupled
to the load, wherein the pass current flows between the second and third
terminals and there is a transconductance relationship between the pass
current and the voltage difference between the first and third terminals;
controlling, in response to the load voltage and an input reference
voltage, the voltage-controlled current device by a control circuit so as
to regulate the load voltage in accordance with the input reference
voltage, the control circuit having an output with an output voltage
coupled to the first terminal of the voltage-controlled current device;
and
limiting the pass current in the voltage-controlled current device by
forcing the voltage-controlled current device into an OFF state upon
determining a first voltage at a first node is less than a second voltage
at a second node, where the first voltage is a first function of the
voltage at the third terminal of the voltage-controlled current device and
the second voltage is a second function of the voltage at the first
terminal of the voltage-controlled current device.
19. The method as set forth in claim 18, further comprising the steps of:
bringing the first voltage to a predetermined voltage when the second
voltage exceeds the first voltage; and
decreasing the second voltage when the first voltage is brought to the
predetermined voltage so that the voltage-controlled current device is OFF
for a length of time during which the second voltage is greater than the
first voltage.
20. The method as set forth in claim 19, wherein:
the first node is the internal node of a first voltage divider with one end
at a voltage equal to a source voltage and another end coupled to the
third terminal of the voltage-controlled current device, and wherein a
first capacitor connects the first node to the third terminal of the
voltage-controlled current device; and
the second node is the internal node of a second voltage divider with one
end grounded and another end at a third node, wherein a second capacitor
connects the third node to the third terminal of the voltage-controlled
current device and a diode connects the third node to the output of the
control circuit.
Description
FIELD OF THE INVENTION
This invention relates to a soft-start switch with a MOSFET. More
particularly, this invention relates to a soft-start switch in which the
voltage drop across the soft-start switch is regulated, the current
supplied to a load is kept below a maximum current value without the need
for a sense resistor by employing the transconductance relationship
between the gate-source voltage and the drain-source current of the
MOSFET, and in which the soft-start function is performed automatically
when a load is applied, without the need of additional sense signals.
BACKGROUND OF THE INVENTION
A soft-start switch is a switching device placed between a power supply and
a load. The soft-start switch when first turned ON provides to the load a
voltage that gradually rises from zero to some desired level. Often the
rise in voltage takes the form of the familiar rising voltage vs. time
curve of a charging capacitor in an RC circuit. See, for example, FIG. 1
where the voltage supplied to the load, denoted as V.sub.out,
exponentially rises to a reference voltage, denoted as V.sub.ref.
It is desirable to add a current limiting feature to a soft-start switch so
that the current supplied to a load is kept below some maximum current
value, so as to prevent excessive current damage to the load and the
connectors, and to reduce unwanted perturbations in other circuits powered
by the power supply powering the soft-switch. For example, a hard-disk
drive when first powered-up is largely a capacitive load, and if it is
powered-up by a simple switch it is possible that an excessively large
current may be drawn by the hard-disk drive.
An example of a prior art soft-start switch 1 is illustrated in FIG. 2,
where MOSFET 10 serves as a voltage-controlled current device with gate 12
coupled to the output of op-amp 20, drain 16 coupled to the input 30 of
the soft-start switch 1, and source 14 coupled to the anode of Schottky
diode 40. Input 30 of soft-start switch 1 is coupled to a power supply
(not shown) with voltage V.sub.0. The output 50 of soft-start switch 1
provides a voltage V.sub.out to load 55. Load 55 may be an active load.
Schottky diode 40 is included to prevent current from being drawn back
into soft-start switch 1 if there is a failure in the power supply, but
otherwise it is not important to the functioning of the soft-start switch.
A reference voltage V.sub.ref, where V.sub.ref <V.sub.0, is provided to
terminal 62 of resistor 60 with resistance R. To node 70 is coupled the
other terminal of resistor 60, the non-inverting input 22 of op-amp 20,
and one terminal of capacitor 90 with capacitance C. The other terminal of
capacitor 90 is grounded. Switching means 80 can ground node 70, thereby
discharging capacitor 90 and grounding the non-inverting input 22 of
op-amp 20. The inverting input 24 of op-amp 20 is coupled to output 50,
thus providing feedback by way of the output of op-amp 20 controlling the
gate voltage of MOSFET 10, thereby controlling the drain-source current
and in turn the voltage V.sub.out applied to load 55. The output voltage
of op-amp 20 is assumed to lie between ground and some voltage V.sub.cc,
where V.sub.cc is sufficient to put MOSFET 10 into or close to saturation.
Without loss of generality we let the ground voltage be zero.
The MOSFET is OFF (V.sub.out =0) when switching means 80 grounds node 70.
Assuming capacitor 90 has been fully discharged, soft-start switch 1
initiates a soft-start power-up when switching means 80 decouples node 70
from ground, thereby allowing capacitor 90 to charge. Thus, the voltage of
non-inverting input 22 is given by V.sub.ref [1-exp(-t/RC)]. Because of
the feedback loop, the op-amp adjusts the gate voltage of MOSFET 10 so
that V.sub.out =V.sub.ref [1-exp(-t/RC)], thus providing the soft-start
capability with V.sub.out given in FIG. 1.
Switching means 80 may perform a current limiting function by switching
MOSFET 10 OFF when too much current is being drawn through the MOSFET and
into the load. FIG. 3 illustrates a prior art soft-start switch with
current limiting. Components in FIG. 3 are referenced by the same numeral
as corresponding identical components in FIG. 2. The soft-start switch of
FIG. 3 is a modification of soft-start switch 1 of FIG. 2 in which a sense
resistor 100 is placed in the current path from MOSFET 10 to load 55. The
voltage drop .DELTA.V across sense resistor 100 is coupled via 102 and 104
to switching means 80. When .DELTA.V is greater than some reference
voltage, indicating that the current is too large, switching means 80
grounds node 70, thereby turning the MOSFET OFF.
It should be appreciated that the prior art soft-start switch of FIGS. 2 or
3 regulates V.sub.out in the sense that the drain-source current of MOSFET
10 is controlled via its gate-source voltage so that V.sub.out is made to
follow the non-inverting voltage of op-amp 20. However, it may be more
desirable to regulate the voltage drop V.sub.0 -V.sub.out rather than the
voltage V.sub.out. For example, more than one power supply may provide
power to a soft-start switch, where one power supply serves as a back-up
for the others. The system may be designed so that one power supply can
handle all the power requirements, but it is desirable that all
functioning power supplies share equally in supplying power to the load.
Unbalanced load sharing may happen when the power supply with the Largest
output voltage supplies most of the current, and thereby most of the power
to the load. To achieve load sharing, the power supplies are built such
that the output voltage of a power supply is gradually lowered when it is
determined that there is unequal load sharing. It is therefore desirable
that V.sub.out also drop gradually in the same amount that V.sub.0 drops
when equal load sharing is sought. Consequently, it is more desirable to
regulate the voltage drop V.sub.0 -V.sub.out than V.sub.out.
Another problem associated with the prior art soft-start switch of FIGS. 2
or 3 arises when a capacitive load is hot-plugged to the soft-start
switch. For example, a hard-disk when first powered-up presents a
capacitive load. It is desirable that a hard-disk drive can be unplugged
from the system and replaced with another hard-disk drive "hot-plugged"
into the system, i.e., the new hard-disk drive is coupled to a soft-start
switch without powering down the system. Hot-plugging a capacitive load
brings V.sub.out momentarily close to zero, thereby increasing the voltage
drop across the drain and source terminals of MOSFET 10 to approximately
V.sub.0. Because of parasitic capacitances between the gate and drain and
between the gate and source inherent in a MOSFET, the sudden increase in
voltage drop across the drain and source terminals induces a sudden
increase in gate-source voltage. Because the MOSFET is a transconductance
device (it is a voltage-controlled current source), this increase in
gate-source voltage results in an undesirable high source-drain current.
Although switching means 80 will eventually turn the MOSFET OFF when a
large current surge is detected, it is more desirable that the MOSFET
never turn ON in the first place. Therefore, it is advantageous that a
soft-start switch with no load connected has the MOSFET turned OFF
(gate-source voltage less than the MOSFET threshold voltage) even though
switching means 80 is not grounding node 70 and capacitor 90 is charged,
and that the switching means keeps the MOSFET OFF even when a capacitive
load is hot-plugged to the soft-start switch.
Yet another problem associated with the prior art switch of FIG. 3 is that
power is dissipated through the sense resistor 100. Although sense
resistors have small resistance, a load may draw several or more amps (for
example a hard-disk drive), and therefore the heat dissipation of sense
resistor 100 must be accounted for. Also, accurate sense resistors add an
additional cost.
Therefore, it is desirable that the prior art soft-start switch of FIGS. 1
or 2 be improved such that the voltage drop V.sub.0 -V.sub.out is
regulated, the MOSFET is held OFF when no load is applied or when a
capacitive load is hot-plugged, and current limiting is accomplished
without a sense resistor. The embodiments of the present invention
described hereinafter accomplish these improvements.
SUMMARY OF THE INVENTION
An advantage of the present invention is a soft-start switch with
regulation of voltage drop across the soft-start switch, i.e., V.sub.0
-V.sub.out, so that load sharing among a plurality of power supplies
coupled to the same soft-start switch is facilitated.
Another advantage of the present invention is a soft-start switch in which
a load may be hot-plugged to the soft-start switch without causing a
current surge.
Another advantage of the present invention is a soft-start switch that
automatically soft-starts a hot-plugged load.
Yet another advantage of the present invention is a soft-start switch with
current limiting without the need for a sense resistor.
In the preferred embodiment of the invention to be disclosed, a MOSFET, an
op-amp, a comparator circuit, diodes, and voltage dividers with capacitors
are employed in combination to effectuate a soft-start switch. The MOSFET
and op-amp are configured as a closed-loop feedback circuit in which the
output of the op-amp is coupled to the gate of the MOSFET and the
inverting input of the op-amp is coupled to the output of the soft-start
switch via a voltage divider. A first RC circuit provides a voltage to the
non-inverting input of the op-amp which can be triggered to gradually rise
from a value close to zero (typically one diode voltage drop above ground)
to some reference voltage. The combination of the first RC circuit and
closed-loop feedback circuit controls the current through the MOSFET such
that the output voltage of the soft-start switch rises gradually from a
value close to zero to the reference voltage when the MOSFET is initially
turned ON. Current limiting means are effectuated by a comparator circuit
and voltage dividers with capacitors. The current limiting means brings
the MOSFET to an OFF state and the non-inverting input of the op-amp close
to zero volts if the op-amp charges a diode-capacitor circuit so that the
voltage drop across its capacitor exceeds a pre-determined reference, and
also, once the current limiting means brings the MOSFET to the OFF state,
the current limiting means allows the soft-start switch to begin a
soft-start power-up after a pre-determined time dependent upon the time
constant of a second RC circuit.
BRIEF DESCRIPTION OF THE DRAWINGS
The accompanying drawings explain the principles of the invention in which:
FIG. 1 illustrates a typical output voltage vs. time curve for when a
soft-start switch begins a soft-start power-up;
FIG. 2 illustrates a prior art soft-start switch;
FIG. 3 illustrates a prior art soft-start switch with prior art current
limiting;
FIG. 4 illustrates an embodiment of the invention; and
FIG. 5 illustrates an embodiment of the invention with additional circuitry
for limiting current when the soft-start switch is in a power-up mode.
DETAILED DESCRIPTION OF THE PREFERRED EMBODIMENTS
FIG. 4 illustrates an embodiment of the invention in which components with
a corresponding component in the previous figures are labeled with the
same reference number. The operation of the circuit in FIG. 4 and how it
achieves the advantages of the invention as outlined in the Summary will
now be explained.
The device labeled 110 is an open-collector comparator with inverting input
112 and non-inverting input 114. Pull-up resistor 116 is coupled to a
voltage V.sub.cc, where V.sub.cc >V.sub.ref. If the voltage at input 114
is greater than the voltage at input 112, then the pull-up resistor 116
with voltage V.sub.cc will bring the voltage at node 118 to V.sub.cc,
thereby reverse biasing diode 120 and allowing capacitor 90 to discharge
so that its terminal closest to the bottom of FIG. 4 is at voltage
V.sub.ref. When the voltage at input 114 is less than the voltage at input
112, the comparator brings the voltage at node 118 to ground, which brings
the cathode of diode 120 to ground and node 70 to one diode voltage drop
above ground, thereby allowing capacitor 90 to charge so that the
potential difference across its plates rises from V.sub.0 -V.sub.ref to
approximately V.sub.0. Note that as capacitor 90 is charging, current is
limited by flowing through resistor 130. Without resistor 130, comparator
110 would not be able to rapidly bring node 70 down to one diode voltage
drop above ground because of the finite current capacity of an
open-collector comparator.
Note that one terminal of capacitor 90 is coupled to a terminal of resistor
60 as in FIGS. 2 and 3, but that the other terminal of capacitor 90 is
coupled to input 30 rather than ground. This configuration brings about
some subtle differences when compared to the prior art switch of FIG. 2 or
3. It should be appreciated that capacitor 90 of FIG. 4 is charging when
the voltage difference between its two terminals is increasing, and is
discharging when the voltage difference is decreasing. For purposes of
explaining the embodiments of the present invention, we shall refer to
capacitor 90 as charged when the voltage difference between its terminals
is approximately V.sub.0 and as discharged when the voltage difference is
V.sub.0 -V.sub.ref. Unlike the prior art switch of FIG. 2 or 3, the RC
circuit in FIG. 4 defined by resistor 60 and capacitor 90 presents to node
70 the voltage V.sub.ref when it is discharged, and presents to node 70
approximately zero volts (one diode voltage drop above ground) when it is
charged. The voltage at node 70 when diode 120 is reverse biased will
still be approximately governed by the equation V.sub.ref [1-exp(-t/RC)]
as for the prior art switch of FIG. 2 or 3, but now t=0 refers to the time
that capacitor 90 starts from a charged state in which the potential
difference across its terminals is approximately V.sub.0 and begins to
discharge to a final potential difference of V.sub.0 -V.sub.ref.
The advantage obtained over the prior art by coupling one terminal of
capacitor 90 to input 30 rather than to ground is that fluctuations in the
voltage V.sub.0 applied to input 30 will cause similar fluctuations in the
voltage at non-inverting input 22, and consequently similar fluctuations
in output voltage V.sub.out by way of the feedback means accomplished by
op-amp 20. This feature is desirable if V.sub.0 is being purposely reduced
because of the load sharing problem as discussed earlier. In other words,
by coupling one terminal of capacitor 90 to input 30 rather than to
ground, the circuit of FIG. 4 is regulating the voltage drop V.sub.0
-V.sub.out, rather than V.sub.out directly, thereby achieving one of the
advantages of the invention.
The soft-start switch of FIG. 4 may be modified in which the terminal of
the capacitor coupled to input 30 is instead coupled to ground, as in the
prior art. Such a modified soft-start switch will achieve the other
advantages of the present invention, but will not have the additional
advantage of regulating the voltage drop V.sub.0 -V.sub.out rather than
V.sub.out directly.
Note that inverting input 24 of op-amp 20 is no longer coupled directly to
output 50 as in the prior art switch of FIG. 2 or 3, but is instead
coupled to node 140 of the voltage divider defined by resistors 142 and
144. The resistance of resistor 142 is chosen substantially larger than
the resistance of resistor 144 so that the voltage at node 140 is close to
V.sub.out when load 55 is present. However, consider the case in which
load 55 is not present, or when it is an infinite impedance, in which case
there is no current flowing through resistors 142 and 144, which brings
the voltage at node 140 to V.sub.0. Then the voltage at inverting input 24
of op-amp 20 is at V.sub.0. However, the voltage at the non-inverting
input 22 is never larger than V.sub.ref, which is lower than V.sub.0, and
therefore the output of op-amp 20 is saturated low at ground.
Consequently, when no load is present, gate 12 of MOSFET 10 is held at
ground even though capacitor 90 may be discharged. Therefore, hot-plugging
a capacitive load, such as a hard-disk drive, will not immediately cause
an increase in gate voltage due to parasitic capacitances within the
MOSFET because the gate 12 is initially held at ground. However,
hot-plugging a capacitive load will quickly bring V.sub.out to zero
momentarily, which will bring the voltage at inverting input 24 close to
zero, in which case the output voltage of op-amp 20 will slew up toward
voltage V.sub.cc which it applies to gate 12. Therefore, to limit current
surge and to initiate a soft-start when a capacitive load is hot-plugged,
it is necessary to continue to keep gate 12 at ground potential and to
bring node 70 to ground potential (or at least within one diode voltage
drop from ground) for at least a period of time sufficiently long so that
capacitor 90 has time to charge. The additional circuitry not yet
discussed in FIG. 4 will achieve these advantages, and will furthermore
provide current limiting if load 55, whether hot-plugged or not, tries to
draw an excessive amount of current. This additional circuitry and its
operation will now be discussed.
Let us continue with the discussion of hot-plugging a capacitive load in
which prior to hot-plugging, the soft-start switch of FIG. 4 is initially
in a state where capacitor 90 is discharged (which assumes that the output
of comparator 110 is V.sub.cc so that diode 120 is reverse biased). As
discussed above, because of the voltage divider defined by resistors 142
and 144, the gate voltage of MOSFET 10 is initially at zero (ground) volts
when no load is present. However, with V.sub.out brought quickly to zero
(ground) due to hot-plugging a capacitive load, the output of op-amp 20
will slew high toward V.sub.cc because the voltage at node 140 will be
close to zero while the voltage at node 70 is still at V.sub.ref. But
because of resistor 145, the series "RC" circuit presented by resistor 145
and the capacitance of gate 12 will charge-up at a slower rate than
capacitor 150 due to the lack of a resistor between capacitor 150 and the
op-amp output (remember that the terminal of capacitor 150 closest to the
top of FIG. 4 is momentarily one Schottky voltage drop above zero volts).
Thus, capacitor 150 will rapidly charge up to toward V.sub.cc when
V.sub.out is brought close to zero due to hot-plugging a capacitive load.
With the voltage at node 160 approaching V.sub.cc, consider the voltage
divider defined by resistors 170a and 170b, which are of equal value. This
voltage divider will present a voltage approaching V.sub.cc /2 at
inverting input 112 of comparator 110. Consider now the voltage divider
defined by resistors 180a and 180b, which are of equal value, and voltage
source 190 with voltage V.sub.lim where V.sub.cc >V.sub.lim (its
significance will be discussed later). The function of capacitor 240 is
discussed later, and for now we ignore its presence when considering the
voltage divider 180a-180b. Consequently, this voltage divider presents a
voltage at non-inverting input 114 close to V.sub.lim /2. Therefore,
because V.sub.cc >V.sub.lim, the output voltage of comparator 110 will go
to zero, which rapidly brings gate 12 and node 70 to one diode voltage
drop above zero because of diodes 200 and 120, respectively. Thus, the
MOSFET stays in the OFF state, thereby keeping V.sub.out at zero and
limiting current to the capacitive load, and capacitor 90 charges.
Furthermore, the ratio of the resistance of resistor 142 to to the
resistance of resistor 144 is chosen such that the voltage at inverting
input 24 will be larger than one diode voltage drop for most practical
values of V.sub.0 and therefore the output of op-amp 20 will saturate to
zero. Also, with the output voltage of comparator 110 at zero, diode 220
is forward biased, and therefore clamps the input 114 to one diode voltage
drop above ground.
We therefore see that hot-plugging a capacitive load puts the soft-start
switch of FIG. 4 in a state where MOSFET 10 is OFF, V.sub.out is zero,
capacitor 90 is charging, the output of op-amp 20 is saturated to zero,
input 114 is at one diode voltage drop above ground, comparator 110 is at
zero volts output, and capacitor 150 is charged up to V.sub.cc. The
soft-start switch of FIG. 4 will now soon be ready to soft-start load 55,
which we now discuss.
With diode 210 now reverse biased (because op-amp 20 is saturated to zero
voltage output), capacitor 150 will now discharge through resistors 170a
and 170b to ground. The voltage at 112 will decay with a time constant
determined by capacitor 150 and resistors 170a and 170b. Eventually the
voltage at 112 will decay below one diode voltage drop, in which case node
118 is pulled up by resistor 116 to a voltage of V.sub.cc, thereby reverse
biasing diodes 120, 200, and 220, and allowing capacitor 90 to discharge
and the soft-start switch to soft-start load 55. The time constant of
capacitor 150 and resistors 170a and 170b should be chosen to be
sufficiently long so that capacitor 90 has time to be fully charged before
a soft-start power-up begins.
Therefore from the above discussion, we see that the soft-start switch of
FIG. 4 achieves the advantage of allowing a capacitive load, such as a
hard-disk drive, to be hot-plugged without a large surge in current and
furthermore provides automatic soft-starting of the hot-plugged load.
Now consider the case in which the circuit of FIG. 4 with load 55 is in a
steady state where capacitor 90 is discharged, node 118 is at voltage
V.sub.cc (i.e., comparator 110 is at output voltage V.sub.cc and diodes
120, 200, and 220 are reversed biased), and MOSFET 10 is ON. We now
discuss how the circuit of FIG. 4 limits current to load 55 if the load
tries to draw an excessive amount of current. For example, the load may be
a hard-disk drive malfunctioning.
First, consider the voltage dividers 170a-170b and 180a-180b. Nodes 230a
and 230b are at the same voltage, which is the source voltage V.sub.s of
source 14 of MOSFET 10. Node 160 is, to within one diode-voltage drop,
equal to the gate voltage V.sub.g of gate 12. (The voltage drop across
resistor 145 can be ignored because of the negligible current drawn by
gate 12.) For simplicity, we ignore the small forward voltage drop across
diode 210. It can easily be shown that the voltage divider 170a-170b
presents a voltage of V.sub.- =V.sub.g /2=(V.sub.s +V.sub.gs)/2 to input
112 where V.sub.gs is the gate-source voltage. Also, it can be shown that
the voltage divider 180a-180b presents a voltage of V.sub.+ =(V.sub.s
+V.sub.lim)/2 to input 114 (remember that the output of the comparator is
at V.sub.cc so that diode 220 is reversed biased). Consequently, the
comparator will change its state from a high voltage of V.sub.cc to zero
voltage when V.sub.- transitions above V.sub.+, or equivalently, when
V.sub.gs transitions above V.sub.lim.
We thus see that the sub-circuit within the dashed lines referenced with
numeral 185 presents to comparator 110 two voltages indicative of whether
V.sub.gs is smaller or greater than V.sub.lim, ignoring the effect of
capacitor 240 on the function of the divider. Other equivalents of
sub-circuit 185 can be constructed by one of ordinary skill in the art of
electronics. The effect of capacitor 240 on the circuit will be discussed
shortly.
By taking advantage of the transconductance associated with MOSFET 10,
sub-circuit 185 will turn MOSFET 10 OFF if load 55 tries to draw an
excessive amount of current. The transconductance of a MOSFET is denoted
by G, where I.sub.D =GV.sub.gs and I.sub.D is the source-drain current. We
assume that the MOSFET is not put into saturation, so that the
transconductance equation holds. We see that V.sub.gs must increase in
order for I.sub.D to increase. Now suppose that load 55 malfunctions and
tries to draw an excessive amount of current, in other words, the
impedance of load 55 suddenly decreases. The MOSFET can be considered a
voltage-controlled current device. A sudden decrease in the impedance of
load 55 does not immediately cause a larger I.sub.D, but rather, the
voltage V.sub.out decreases. Because of the closed-loop feedback, op-amp
20 will try to keep V.sub.out close to V.sub.ref by increasing its output
voltage so as to increase the gate-source voltage V.sub.gs which in turn
would increase I.sub.D which in turn would increase V.sub.out. In
particular, when the MOSFET is close to saturation, G decreases, so that
an even larger increase in V.sub.gs is required to increase I.sub.D
compared to the case in which the MOSFET is not close to saturation. As
the op-amp tries to increase I.sub.D by increasing V.sub.gs, capacitor 150
is charging up and the voltage presented by voltage divider 170a-170b to
input 112 increases. As discussed above, the comparator will go into the
zero voltage output state when V.sub.gs transitions above V.sub.lim.
Consequently, the value of V.sub.lim determines the maximum drain-source
current, I.sub.D (max), that the soft-start switch circuit of FIG. 4 will
allow, where I.sub.D (max)=GV.sub.lim.
Thus, if the gate-source voltage V.sub.gs transitions above V.sub.lim, we
have the situation discussed earlier in which the MOSFET is driven OFF,
capacitor 90 begins to charge, and diode 220 brings the voltage at input
114 to one diode voltage drop above ground. The soft-start switch will
then begin a soft-start power-up once the voltage at input 112 decays to a
value less than one diode voltage drop. The utility of diode 220 is now
clear. It provides positive feedback, so that just after the voltage at
input 112 transitions above the voltage at input 114, it brings the
voltage at 114 close to ground so that the time interval needed for the
voltage at input 112 to decay below the voltage at input 114 is sufficient
for capacitor 90 to be fully charged.
Therefore, the soft-start switch of FIG. 4 limits current through load 55
by turning MOSFET 10 OFF and beginning a soft-start. Consequently, if load
55 is permanently malfunctioning, the soft-start switch of FIG. 4 will
repeatedly go through shut-down and soft-start cycling until the
malfunctioning load is removed. In the case in which load 55 is a
hard-disk drive, a soft-start switch undergoing shut-down and soft-start
cycling indicates that the hard-disk drive it powers is malfunctioning and
that therefore the system operator can remove the hard-disk drive and
hot-plug a new hard-disk drive.
It should be appreciated that the soft-start switch circuit of FIG. 4
accomplishes current limiting without the need of a sense resistor. The
power dissipated by the voltage dividers 142-144, 170a-170b, and 180a-180b
can be made very small by choosing large values for the resistances. In
practice, for driving hard-disk drives, the current through these voltage
dividers is on the order of milliamps whereas the drain-source current
I.sub.D is on the order of amps.
We now consider the effect of capacitor 240 in the circuit of FIG. 4.
Capacitor 240 feeds-forward changes in V.sub.out to input 114 of
comparator 110. If V.sub.out is changing slowly relative to the time
constant of capacitor 240 and resistors 180a and 180b, capacitor 240 does
not affect the voltage at comparator input 114. However, if V.sub.out is
chancing quickly relatively to the time constant of capacitor 240 and
resistors 180a and 180b, then it will affect input 114. Of primary
importance is the case when V.sub.out is decreasing quickly, as would be
the case during an initial hot-plugging of a capacitive load, or if a load
were to fail and short the output 50 of the soft-start switch to ground.
In this case, capacitor 240 would force the voltage at input 114 to be
temporarily lower than it would otherwise be if capacitor 240 were not
present. This action effectively lowers the trip threshold of comparator
110 and makes it easier for comparator 110 to turn MOSFET 10 OFF. In fact,
for large and fast changes in V.sub.out, comparator 110 shuts down MOSFET
10 immediately, without waiting for the voltage at node 160 to increase.
Thus we see that capacitor 240 aids the soft-start switch in shutting down
quickly during an initial hot-plugging of a load. Also, we see that
capacitor 240 provides for shut-down of the soft-start switch of FIG. 4
when there is an instantaneous short in load 55 after the soft-start
switch has already soft-started load 55.
Capacitors 250 and 260 add additional phase margin to the control loop of
the op-amp so that the control loop is stable. Capacitor 270 filters load
generated noise in the output voltage of the soft-switch. Capacitors 250,
260, and 270 are not directly relevant to the scope of the present
invention, but are included in FIG. 4 because they would be included in a
preferred embodiment.
An additional transistor and resistor may be added to the circuit as shown
in FIG. 5, where in this figure we have only shown the additional
components and Schottky diode 40 and MOSFET 10 of FIG. 4. Not shown in
FIG. 5 are the remaining components of FIG. 4, which are assumed to be
incorporated into FIG. 5. The additional circuitry shown in FIG. 5 is
desirable for the following reason. When MOSFET 10 is not near saturation,
the transconductance G is larger than for the case when MOSFET 10 is near
saturation. Therefore, if a fault in load 55 should occur while the MOSFET
is not near saturation, for example when the soft-start switch is in the
soft-start power-up mode, then V.sub.lim may be set too high for this
larger transconductance case and consequently too much drain-source
current I.sub.D may be allowed to flow through the MOSFET and into the
load. The additional circuitry shown in FIG. 5 can solve this problem
depending upon the choice of resistor 290. When an excessive current is
drawn through Schottky diode 40, its voltage drop increases, which can
bring transistor 280 into conduction, thereby decreasing the voltage of
gate 12 and limiting the MOSFET conduction. This effectively opens the
control loop and results in the output of op-amp 20 to slew toward
V.sub.cc, resulting in a shutdown as previously described.
Table 1 provides an example of nominal values for the resistors,
capacitors, and voltages in the embodiment of FIGS. 1 and 2 for the case
in which the load is a hard-disk drive. Other values may be used.
Numerous modifications may be made to the embodiments described above
without departing from the spirit and scope of the invention. For example,
it was already discussed that an operable soft-start switch would arise
from modifying FIG. 4 in which the terminal of capacitor 90 coupled to
input 30 is instead coupled to ground. As another example, FIG. 4 may be
modified in which the inverting input 24 of op-amp 20 is coupled directly
to output 50 rather than through the voltage divider 142-144. For yet
another example, comparator 110 need not be coupled to gate 14 via diode
200. Although such modifications would lead to operable soft-start
switches, they are not preferable to the embodiment of FIG. 4 because they
would lack some advantages. However, such modifications of FIG. 4, and
others, would still result in soft-start switches which employ the
transconductance of MOSFET 10 without the need for a current sense
resistor. Also, other voltage-controlled current devices other than a
MOSFET may be substituted.
TABLE I
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resistor 60 487K.OMEGA.
resistor 130 2K.OMEGA.
capacitor 90 22000pF
resistor 142 10K.OMEGA.
resistor 144 1000.OMEGA.
capacitor 250 15000pF
capacitor 260 15000pF
resistor 145 10K.OMEGA.
capacitor 150 100000pF
capacitor 270 1000pF
resistor 116 100K.OMEGA.
resistors 170a and 170b
487K.OMEGA.
resistors 180a and 180b
100K.OMEGA.
capacitor 240 330pF
V.sub.O 12.8v
V.sub.ref 12V
V.sub.CC 20V
V.sub.lim
5.5V
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