Back to EveryPatent.com
United States Patent |
5,309,344
|
Smith
|
May 3, 1994
|
Dual active clamp, zero voltage switching power converter
Abstract
A dual active clamp (DAC) converter comprising two active-clamp,
zero-voltage switching (ZVS) converters for converting electrical power
from an input source to an output load is described. The two active-clamp
ZVS converters are coupled in parallel to one another at their respective
inputs and outputs, and are operated in a manner which reduces the overall
input and output ripple currents of the DAC converter, which reduces the
EMI emissions of the DAC converter. Each of the active-clamp ZVS
converters achieves zero-voltage switching on its primary switch means,
which also improves EMI emissions. Each active-clamp ZVS converter
includes a transformer having a primary winding and a secondary winding,
the secondary winding being coupled to the output load and the primary
switch means being coupled between the primary winding and a voltage
source, and an active clamp circuit comprising a series combination of a
storage capacitor and a first switch means. The active clamp is coupled to
the transformer to capture energy released from the transformer at
specific times, which is then used to reset the transformer's core and to
create a zero voltage switching condition across the primary switch means.
A second switch means is connected in series with the secondary winding of
the transformer and is operated to prevent the loading effects of the
secondary from interrupting the creation of the zero-voltage switching
condition.
Inventors:
|
Smith; David A. (Kowloon, HK)
|
Assignee:
|
Astec International, Ltd. (Hong Kong, HK)
|
Appl. No.:
|
943008 |
Filed:
|
September 9, 1992 |
Current U.S. Class: |
363/20; 363/65; 363/71; 363/97; 363/131 |
Intern'l Class: |
H02M 003/335 |
Field of Search: |
363/20,21,65,71,86,89,95,97,131
|
References Cited
U.S. Patent Documents
4417153 | Nov., 1983 | Onodera et al. | 363/20.
|
4441146 | Apr., 1984 | Vianciarelli | 363/20.
|
4559590 | Dec., 1985 | Davidson | 363/21.
|
4618919 | Oct., 1986 | Martin | 363/21.
|
4635179 | Jan., 1987 | Carsten | 363/70.
|
4669039 | May., 1987 | Brakus | 363/71.
|
4809148 | Feb., 1989 | Barn | 363/131.
|
4870554 | Sep., 1989 | Smith | 363/131.
|
4870555 | Sep., 1989 | White | 363/21.
|
4959764 | Sep., 1990 | Bassett | 363/20.
|
4975821 | Dec., 1990 | Lethallier | 363/97.
|
5001620 | Mar., 1991 | Smith | 363/89.
|
5126931 | Jun., 1992 | Jitaru | 363/21.
|
5173846 | Dec., 1992 | Smith | 363/20.
|
5229928 | Jul., 1993 | Karlsson et al. | 363/65.
|
Foreign Patent Documents |
60-96175 (A) | May., 1985 | JP | .
|
Other References
Carsten, "High Power SMPS Require Intrinsic Reliability," Proceedings of
the PCI, Mar. 1982, pp. 456-471.
Carsten, "Design Tricks, Techniques and Tribulations at High Conversion
Frequencies," Proceedings of the HPFC, Apr. 1987, pp. 139-152.
UNITRODE Integrated Circuits Corp., datasheet for the "UC1825 High Speed
PWM Controller," Linear Integrated Circuits Data and Applications
Handbook, Apr. 1990, pp. 4-120 through 4-126.
ASTEC Semiconductor, datasheet for the "ASTEC AS3842/3/4/5 Current Mode PWM
Controller," Sep. 1990, pp. 1-12.
UNITRODE Integrated Circuits Corp., datasheet for the "UC1842/3/4/5 Current
Mode PWM Controller," Linear Integrated Circuits Databook, Dec. 1987, pp.
3-107 to 3-112.
UNITRODE Inegrated Circuits Corp., application notes for "UC3842/3/4/5
Provides Low-Cost Current-Mode Control" and UC3842/3/4/5 Series of
Current-Mode PWM IC's, Linear Integrated Circuits Databook, Dec. 1987, pp.
9-105 to 9-118.
|
Primary Examiner: Sterrett; Jeffrey
Attorney, Agent or Firm: McCubbrey, Bartels, & Ward
Parent Case Text
This application is a continuation-in-part of U.S. patent application Ser.
No. 07/668,587, filed Mar. 13, 1991 for Zero Voltage Switching Power
Converter now U.S. Pat. No. 5,173,846.
Claims
What is claimed is:
1. A dual active clamp forward converter comprising:
an input port for receiving a source of power and an output port for
providing power to a load;
a first converter and a second converter, each said converter having a
power transformer including a primary winding coupled to said input port
and a secondary winding coupled to said output port, a primary switch
means coupled in series between said primary winding of said power
transformer and said input port, a series combination of a first switch
means and a first capacitor, said series combination of said first switch
means and said first capacitor coupled in parallel with one of said
primary winding, said secondary winding, and said primary switch means,
and a second switch means coupled in series with said secondary winding of
said power transformer, said primary switch means when closed causing a
first portion of energy from the source of power at said input port to be
coupled through said power transformer to the load by means of said
secondary winding and causing a second portion of said energy to be stored
in said power transformer;
means for generating a first duty-cycle signal and a second duty-cycle
signal, each of said first and second duty-cycle signals having a first
state and a second state and operative to alternate in time between said
first and second states, said second duty-cycle signal being phase-shifted
in time by substantially 180 degrees from said first duty-cycle signal;
means responsive to said first duty-cycle signal for operating the primary
switch means and first switch means of said first converter, said means
causing the primary switch means of said first converter to close
substantially when said first duty-cycle signal enters its respective
first state and to open substantially when said first duty-cycle signal
enters its respective second state, said means causing the first switch
means of said first converter to open substantially when said first
duty-cycle signal enters its respective first state and to close
substantially when said first duty-cycle signal enters its respective
second state, said means opening the first switch means of said first
converter prior to when the primary switch means of said first converter
closes to create a zero-voltage switching condition on said primary switch
means when said primary switch means next is caused to close;
means for operating the second switch means of said first converter, said
means opening the second switch means of said first converter
substantially when the primary switch means of said first converter opens
and closing said second switch means substantially when said primary
switch means closes, said second switch means being operated such that
substantially no current is conducted by said secondary winding of said
first converter during the interval after said first switch means is
opened and before said primary switch means is closed;
means responsive to said second duty-cycle signal for operating the primary
switch means and the first switch means of said second converter, said
second control means causing the primary switch means of said second
converter to close substantially when said second duty-cycle signal enters
its respective first state and to open substantially when said second
duty-cycle signal enters its respective second state, said means causing
the first switch means of said second converter to open substantially when
said second duty-cycle signal enters its respective first state and to
close substantially when said second duty-cycle signal enters its
respective second state, said means opening the first switch means of said
second converter prior to when the primary switch means of said second
converter closes to create a zero-voltage switching condition on said
primary switch means when said primary switch means next is caused to
close; and
means for operating the second switch means of said second converter, said
means opening the second switch means of said second converter
substantially when the primary switch means of said second converter opens
and closing said second switch means substantially when said primary
switch means, said second switch means being operated such that
substantially no current is conducted by the secondary winding of said
second converter during the interval after said first switch means is
opened and before said primary switch means is closed.
2. The dual active clamp forward converter of claim 1 wherein each of said
first and second duty-cycle signals further comprises a plurality of
consecutive switching periods, each said switching period having a
substantially constant time duration T, each said switching period
beginning when its corresponding duty-cycle signal enters its respective
first state from its respective second state, continuing when its
corresponding duty-cycle signal enters its respective second state from
its respective first state, and ending when its corresponding duty-cycle
signal next enters its respective first state from its respective second
state.
3. The dual active clamp forward converter of claim 2 wherein the
consecutive switching periods of said second duty-cycle signal start
substantially one-half of said time duration T after the consecutive
switching periods of said first duty-cycle signal start.
4. The dual active clamp forward converter of claim 2 wherein the ratio of
the duration said first duty-cycle signal is in its said first state to
the duration of said time duration T is between 0.30 and 0.70, and wherein
the ratio of the duration said second duty-cycle signal in its said first
state to the duration of said time duration T is between 0.30 and 0.70.
5. The dual active clamp forward converter of claim 2 wherein the ratio of
the duration said first duty-cycle signal is in its said first state to
the duration of said time duration T is between 0.40 and 0.65, and wherein
the ratio of the duration said second duty-cycle signal in its said first
state to the duration of said time duration T is between 0.40 and 0.65.
6. The dual active clamp forward converter of claim 2 wherein the ratio of
the duration said first duty-cycle signal is in its said first state to
the duration of said time duration T is substantially the same as the
ratio of the duration said second duty-cycle signal in its said first
state to the duration of said time duration T.
7. The dual active clamp forward converter of claim 2 wherein said means
for generating said first and second duty cycle signals comprises means
for regulating the output voltage at said output port within a range
around a predetermined target value, wherein the voltage at said input
port is at a nominal value when the voltage at said output port is
substantially at said predetermined target value, and wherein the ratio of
the duration said first duty-cycle signal is in its said first state to
the duration of said time duration T is substantially 0.50 and the ratio
of the duration said second duty-cycle signal in its said first state to
the duration of said time duration T is substantially 0.50 when said input
voltage is at its nominal value and the voltage at said output port is at
said predetermined target value.
8. The dual active clamp forward converter of claim 1 wherein said means
for operating the second switch means of said first converter opens said
second switch means after the primary switch means of said first converter
opens and before the first switch means of said first converter is closed.
9. The dual active clamp forward converter of claim 8 wherein said means
for operating the second switch means of said second converter opens said
second switch means of said second converter after the primary switch
means of said second converter opens and before the first switch means of
said second converter is closed.
10. The dual active clamp forward converter of claim 1 wherein said means
for operating the second switch means of said first converter closes said
second switch means after the primary switch means of said first converter
closes such that the primary switch means of said first converter become
completely conductive before it is caused to conduct the current of the
secondary winding of said first converter which is reflected to the
primary winding of said first converter by transformer action and thereby
to said primary switch means.
11. The dual active clamp forward converter of claim 10 wherein said means
for operating the second switch means of said second converter closes the
second switch means of said second converter after the primary switch
means of said second converter closes such that the primary switch means
of said second converter become completely conductive before it is caused
to conduct the current of the secondary winding of said second converter
which is reflected to the primary winding of said second converter by
transformer action and thereby to said primary switch means.
12. The dual active clamp forward converter of claim 1 wherein the primary
switch means of said first converter comprises:
a switch connected in series between the primary winding of said first
converter and said input port, said switch responsive to said first
control means; and
means coupled in parallel with said switch for detecting a zero voltage
switching condition across the primary switch means of said first
converter and for conducting current upon the detection of said zero
voltage switching condition across the primary switch means of said first
converter.
13. The dual active clamp forward converter of claim 12 wherein said input
port includes a positive terminal and a negative terminal, a more positive
voltage being applied to said positive terminal by the source of power,
and wherein said means for detecting a zero voltage switching condition
across the primary switch means of said first converter comprises a
rectifier having an anode terminal and a cathode terminal, said rectifier
connected in parallel with said switch of the primary switch means such
that said rectifier conducts current directed toward said positive
terminal of said input port.
14. The dual active clamp forward converter of claim 1 wherein the series
combination of the first switch means and the first capacitor of said
first converter is coupled in parallel with the primary winding of said
first converter.
15. The dual active clamp forward converter of claim 1 wherein the series
combination of the first switch means and the first capacitor of said
first converter is coupled in parallel with the secondary winding of said
first converter.
16. The dual active clamp forward converter of claim 1 wherein the series
combination of the first switch means and the first capacitor of said
first converter is coupled in parallel with the primary switch means of
said first converter.
17. The dual active clamp forward converter of claim 1 wherein said dual
active clamp converter further comprises a second capacitor coupled in
parallel with said output port, and wherein each said converter further
comprises:
a choke inductor coupled in series with said second capacitor, the series
combination of said choke inductor and said second capacitor being coupled
in parallel with the series combination of the second switch means of said
converter and the secondary winding of said converter; and
a rectifier coupled in parallel with said series combination of said choke
inductor and said second capacitor, said rectifier for coupling current
from said choke inductor to said output port when the second switch means
of said converter is open.
18. The dual active clamp forward converter of claim 1 wherein said means
for operating the primary switch means and the first switch means of said
first converter comprises means for generating a first control signal for
controlling the operation of the primary switch means of said first
converter and a second control signal for controlling the operation of the
first switch means of said first converter, wherein said means for
operating the second switch means of said first converter comprises means
for generating a third control signal for controlling the operation of the
second switch means of said first converter, each of said control signals
having a first state for directing its respective switch to close and a
second state for directing its respective switch to open, wherein said
first and second control signals are generated such that only one of said
first and second control signals is in its respective said first state at
any given time, and wherein said third control signal enters its said
first state a predetermined time duration after said first control signal
enters its said first state and enters its said second state a
predetermined time duration after said first control signal enters its
said second state.
19. The dual active forward converter of claim 1 wherein said means for
operating the second switch means of said first converter comprises:
detection means for detecting the voltage across the secondary winding of
said first converter and for generating a first signal having a first
state when the voltage across said secondary winding is positive and a
second state when the voltage across said secondary winding is negative;
delay generation means responsive to said first signal for generating a
switch control signal to control the second switch means of said first
converter, said switch control signal having a first state for directing
said second switch means to close and a second state for directing said
second switch means to open, said switch control signal entering said
first state a first predetermined time duration after said first signal
enters its first state and entering said second state a second
predetermined time duration after said first signal enters its second
state; and
coupling means for coupling said switch control signal to the second switch
means of said first converter.
20. The dual active claim forward converter of claim 19 wherein said first
predetermined time duration is longer than said second predetermined time
duration.
21. The dual active clamp forward converter of claim 19 wherein said
detection means comprises a series combination of a resistor and a Zener
diode, said Zener diode having an anode terminal and a cathode terminal,
said resistor having a first terminal and a second terminal coupled to
said cathode terminal of said Zener diode, said series combination coupled
across the secondary winding of said first converter such that a positive
voltage is produced at said cathode terminal of said Zener diode when said
secondary winding provides current to said load, said first signal being
provided at said cathode terminal of said Zener diode.
22. The dual active clamp forward converter of claim 21 wherein said
detection means further comprises a rectifier coupled in parallel with
said Zener diode, said rectifier having a anode terminal connected to said
anode terminal of said Zener diode and a cathode terminal connected to the
cathode terminal of said Zener diode.
23. The dual active clamp forward converter of claim 19 wherein said delay
generation means comprises:
a first inverter having an input coupled to said first signal and an
output, said output having a logic high state and a logic low state;
a second inverter having an input coupled to said output of said first
inverter and an output for providing said switch control signal for the
second switch means of said first converter; and
delay means for introducing a first signal propagation delay between said
first and second inverters when said output of said first inverter changes
from its said logic low state to its said logic high state and for
introducing a second signal propagation delay between said first and
second inverters when said output of said first inverter changes from its
said logic high state to its said logic low state, said first and second
signal propagation delays such that said switch control signal enters its
said first state said first predetermined time duration after said first
signal enters its first state and said switch control signal enters its
said second state said second predetermined time duration after said first
signal enters its second state.
24. The dual active clamp forward converter of claim 23 wherein said delay
means comprises:
a rectifier having an anode terminal connected to said output of said first
inverter and a cathode terminal connected to said input of said second
inverter;
a capacitor having a first terminal connected to said input of said second
inverter and a second terminal coupled to a fixed potential reference; and
a resistor having a first terminal connected to said input of said second
inverter and a second terminal coupled to said fixed potential reference.
25. The dual active clamp forward converter of claim 19 wherein the second
switch means of said first converter comprises a field effect transistor
having a gate terminal, a drain terminal connected to the secondary
winding of said first converter, and a source terminal coupled to said
output port, and wherein said coupling means for coupling said control
signal to said second switch means of said first converter comprises a
coupling transformer having a first winding coupled to said control signal
and a second winding having a first terminal coupled to said gate terminal
of said second switch means and a second terminal coupled to said source
terminal of said second switch means.
26. The dual active clamp forward converter of claim 25 wherein said
coupling means for coupling said control signal to said second switch
means further comprises a resistor coupled in series between said gate
terminal and said second terminal of said second winding.
27. A dual active clamp forward converter comprising:
an input port for receiving a source of power and an output port for
providing power to a load;
a first converter and a second converter, each said converter having a
power transformer including a primary winding coupled to said input port
and a secondary winding coupled to said output port, a primary switch
means coupled in series between said primary winding of said power
transformer and said input port, a series combination of a first switch
means and a first capacitor, said series combination of said first switch
means and said first capacitor coupled in parallel with one of said
primary winding, said secondary winding, and said primary switch means,
and a second switch means coupled in series with said secondary winding of
said power transformer, said primary switch means when closed causing a
first portion of energy from the source of power at said input port to be
coupled through said power transformer to the load by means of said
secondary winding and causing a second portion of said energy to be stored
in said power transformer;
means for generating a first duty-cycle signal and a second duty-cycle
signal, each of said first and second duty-cycle signals having a first
state and a second state and operative to alternate in time between said
first and second states, said second duty-cycle signal being phase-shifted
in time by substantially 180 degrees from said first duty-cycle signal;
a first control means responsive to said first duty-cycle signal for
operating the primary switch means, the first switch means, and the second
switch means of said first converter, said first control means causing the
primary switch means of said first converter to close substantially when
said first duty-cycle signal enters its respective first state and to open
substantially when said first duty-cycle signal enters its respective
second state, said first control means causing the first switch means of
said first converter to open substantially when said first duty-cycle
signal enters its respective first state and to close substantially when
said first duty-cycle signal enters its respective second state, said
first control means opening the first switch means of said first converter
prior to when the primary switch means of said first converter closes to
create a zero-voltage switching condition on said primary switch means
when said primary switch means next is caused to close, said first control
means opening the second switch means of said first converter
substantially when the primary switch means of said first converter opens
and closing said second switch means substantially when said primary
switch means closes, said second switch means being operated such that
substantially no current is conducted by said secondary winding of said
first converter during the interval after said first switch means is
opened and before said primary switch means is closed; and
a second control means responsive to said second duty-cycle signal for
operating the primary switch means, the first switch means, and the second
switch means of said second converter, said second control means causing
the primary switch means of said second converter to close substantially
when said second duty-cycle signal enters its respective first state and
to open substantially when said second duty-cycle signal enters its
respective second state, said second control means causing the first
switch means of said second converter to open substantially when said
second duty-cycle signal enters its respective first state and to close
substantially when said second duty-cycle signal enters its respective
second state, said second control means opening the first switch means of
said second converter prior to when the primary switch means of said
second converter closes to create a zero-voltage switching condition on
said primary switch means when said primary switch means next is caused to
close, said second control means opening the second switch means of said
second converter substantially when the primary switch means of said
second converter opens and closing said second switch means substantially
when said primary switch means, said second switch means being operated
such that substantially no current is conducted by the secondary winding
of said second converter during the interval after said first switch means
is opened and before said primary switch means is closed.
28. The dual active clamp converter of claim 27 wherein said first control
means opens the second switch means of said first converter after the
primary switch means of said first converter opens and before the first
switch means of said first converter is closed.
29. The dual active clamp converter of claim 28 wherein said second control
means opens the second switch means of said second converter after the
primary switch means of said second converter opens and before the first
switch means of said second converter is closed.
Description
FIELD OF THE INVENTION
The present invention relates to DC-to-DC switching power converters which
transfer power from a source supply at a given voltage potential to a
destination load at a different voltage potential. In particular, the
invention relates to the control of energy in transformers of DC-to-DC
forward converters.
BACKGROUND OF THE INVENTION
The forward converter is a common circuit topology used to transform
electric energy from a source at a given potential to a destination load
at a different potential. Typically, the forward converter requires fewer
components than other converter topologies and is smaller and lighter. The
standard forward converter comprises a transformer having a primary
winding, a secondary winding, and a third winding. The primary winding is
coupled to a source of power, usually DC power, via a primary switch and
the secondary winding is coupled to a load via two commutating diodes. The
primary switch generally comprises a semiconductor switching device such
as a field-effect transistor (FET) or a bipolar-junction transistor (BJT).
When the primary winding is energized by the closing of the primary
switch, energy is immediately transferred to the secondary winding, hence
the name forward converter. The third winding is coupled to the power
source via a rectifying diode and serves to reset the ferromagnetic core
of the transformer when the primary switch is opened. The transformer's
third winding provides a current path for discharging the transformer's
magnetizing current, which is created when the primary winding is
energized.
The standard forward converter is particularly well suited for low and
medium power-conversion applications. However, it is not as efficient as
other topologies in converting power, particularly in low power
applications. The following factors contribute to the lower efficiency of
the forward converter.
First, the core-reset operation in the forward converter using the third
winding is not as efficient as other core-reset operations in other
converter topologies. The rectifier in series with the third winding
causes an amount of power dissipation and, more importantly, the addition
of a third winding to the transformer increases the transformer's
dimensions and thereby reduces the transformer efficiency by increasing
the core losses, the resistance of each winding, and the self inductance
of each winding.
Second, the forward converter only uses the first quadrant of the
transformer's core B-H characteristic, leaving the third quadrant unused.
As a result, the size of the transformer must be increased to enlarge the
operating range of the first quadrant at the expense of higher core losses
and higher winding resistances.
Third, the power dissipation in the primary switch when it is switched on
(a turn-on event) is greater than the power dissipation in comparable
switches in other topologies. The power dissipation in a switch during a
switching event depends upon the product of the voltage across the switch
and the current through the switch. In a forward converter, the voltage
difference across the primary switch changes from a value equal to the
input voltage of the power source to a value near zero when the switch is
closed. Additionally, the primary current begins immediately since the
forward converter provides current to the secondary winding immediately
upon energizing the primary winding. The high input voltage and the
instantaneous current flow in the primary switch leads to a high power
dissipation loss in the primary switch. The power dissipation losses
during switching events become more significant as the switching frequency
of the forward converter is increased, as is done to improve the
conversion efficiency of the converter's transformer. The direct power
dissipation losses become more significant as the switching frequency
increases because the duration of each switching event comprises a larger
fraction of each switching cycle duration as the switching frequency
increases.
The above first and second factors have been addressed in U.S. Pat. No.
4,441,146 issued to Vinciarelli. In U.S. Pat. No. 4,441,146, the third
winding is eliminated and replaced by a series combination of a storage
capacitor and an auxiliary switch coupled across either the primary or
secondary winding. The auxiliary switch is operated counter to the primary
switch, i.e., the auxiliary switch is open when the primary switch is
closed and closed when the primary switch is open. When the primary switch
is open, the storage capacitor and auxiliary switch operate to capture and
store the transformer's magnetizing current, which was built up when the
primary winding was energized, and then to return the magnetizing current
to the transformer in a manner which resets the transformer's core. The
elimination of the third winding addresses the first above efficiency
factor. Additionally, the returning of the magnetizing current to the
transformer creates a condition where the first and third quadrants of the
transformer core's B-H characteristic are utilized, thereby addressing the
second above efficiency factor. This converter is often referred to as an
active clamp forward converter because the series combination of the
auxiliary switch and storage capacitor acts as a voltage clamp which is
actively (i.e., selectively) coupled to the transformer's windings.
The third efficiency factor, however, is not addressed by the prior art. In
this respect, the present invention provides improvements to the basic
forward-converter topology by reducing the power dissipation during
switching events, thereby increasing the power-conversion efficiency of
the basic forward-converter topology.
Another drawback of the standard forward converter, as well as the
active-clamp forward converter, is the problem of excessive "switching
noise" on its input current line. This noise on the input current is
primarily due to the "chopping" of the primary circuit current. The ripple
noise has thus far prevented the use of the standard forward converter and
active clamp forward converter in low noise applications, such as 48
VDC-to-5 VDC supplies for telephone exchange equipment. These ripple
currents tend to create high-frequency electromagnetic interference (EMI),
which can disturb the operation of nearby digital and communication
equipment. Many countries are increasingly regulating the amount of EMI
that can be generated by power supplies, forcing power supply manufactures
to seek more complex, and often more expensive, power supply designs. The
forward converter and active-clamp forward converter arts have attempted
to correct the switching noise problem by incorporating large filtering
capacitors at the inputs of these converters. However, this greatly
increases the size of the converters, and somewhat decreases the
efficiency of the converters since a large ripple current is caused to
flow through the equivalent series resistance (esr) of the input filter
capacitors. The present invention is also directed towards reducing this
switching noise, and to do so while maintaining a compact converter size
and high efficiency.
Presently, there is a great need for a compact, low noise, and highly
efficient power converter, which is currently not being provided by the
forward converter and active clamp forward converter topologies due to the
above-described problems.
SUMMARY OF THE INVENTION
The present invention recognizes that the magnetizing current in the
forward converter's transformer is a source of energy that could be used
to reduce the voltage across the converter's primary switch to a value
near zero volts before the primary switch is closed to energize the
primary winding. A reduction of voltage to near zero volts in magnitude
(either positive or negative in value) just prior when the primary switch
is closed greatly reduces the power dissipation in the primary switch
means and is referred to as a zero-voltage-switching condition. The
present invention further recognizes that the loading on the converter's
secondary winding diverts some or all of the magnetizing current away from
the primary winding into the secondary winding in the portion of the
converter's switching cycle prior to the closing of the primary switch,
thereby preventing such a zero-voltage-switching condition for the
converter's primary switch under most operating conditions. The present
invention also recognizes that a switch coupled in series with the
secondary winding and operated in a specific manner may be used to block
this loading by the secondary circuit. The present invention further
recognizes that the input ripple current of one active clamp forward
converter has a characteristic waveform which could be substantially
"canceled" by another similar active clamp forward converter operating at
the same switching-frequency but in opposite phase, i.e., at a phase
difference of substantially 180 degrees. The present invention further
recognizes that two such active clamp forward converters operated in this
manner may replace a single active clamp converter to provide the same
output power level, while using substantially the same power transformer
core volume and reducing the size of the input filter capacitance.
Broadly stated, the present invention encompasses a dual active clamp (DAC)
forward converter including two zero-voltage-switching (ZVS) active-clamp
forward converters coupled in parallel and operated in a manner so as to
substantially reduce the input ripple current of the DAC converter so that
the resultant compact size and low cost of the active clamp forward
converters may be exploited for low-noise and high efficiency power supply
applications. The present invention further encompasses, in each of the
two ZVS active-clamp forward converters, means for isolating the loading
effects of the secondary winding from the converter's transformer during
the portion of the converter's switching cycle prior to the closing of the
primary switch and means for reducing the voltage across the converter's
primary switch to a value near zero before the primary switch is closed to
energize the primary winding.
More specifically, the present invention comprises an input port for
receiving a source of power, an output port for providing power to a load,
a first ZVS active-clamp forward converter, and a second ZVS active-clamp
forward converter. Each ZVS active clamp converter includes a power
transformer having a primary winding coupled to the input port and a
secondary winding coupled to the output port, a primary switch means
coupled in series between the primary winding and the input port, and an
active clamp circuit including a series combination of a first switch and
a storage capacitor. In a preferred embodiment of the present invention,
the active clamp circuit in each converter may be coupled across the
primary winding, the secondary winding, or the primary switch means of the
ZVS active-clamp converter. The primary switch means causes energy to be
stored in the transformer when the primary switch means is closed during
an ON-period and causes the energy to be released from the transformer
when the primary switch means is open during an OFF-period.
The present invention further comprises means for generating a first
duty-cycle signal and a second duty-cycle signal for controlling the
switching state of the primary switch means of the respective ZVS
active-clamp forward converters, each of the first and second duty-cycle
signals having a first state and a second state. Each of the first and
second duty-cycle signals alternate between its respective first state and
its respective second state. Additionally, the second duty-cycle signal is
phase-shifted in time by substantially 180 degrees from the first
duty-cycle signal.
For each ZVS active-clamp converter, the present invention comprises
circuitry for isolating the loading effects of the secondary winding from
the converter's transformer in a time interval preceding the closing of
the primary switch means and for utilizing the magnetizing energy stored
in the transformer to reduce the voltage across the primary switch means
to a value near zero before the primary switch means is closed. This
circuitry comprises the above-described active clamp, a second switch
means 150 connected in series with the secondary winding of the
converter's transformer, and a switch control means for operating the
first and second switch means in relation to the ON-periods and
OFF-periods of the primary switch means. The switch control means closes
the first switch substantially when the primary switch means opens and
opens the first switch means prior to when the primary switch means closes
to create a zero-voltage switching condition across the primary switch
means when the primary switch means is next closed. Additionally, the
switch control means closes the second switch means on or after the
primary switch means closes and opens the second switch means before the
first switch means closes. As such, the second switch means remains open
in the portion of the converter's switching cycle prior to the closing of
the primary switch means. This isolates the loading effects of the
secondary winding from the converter's transformer and creates a
zero-voltage-switching across the primary switch means when the primary
switch means is next closed.
The coupling and operation of the first and second converters in the above
described manner significantly reduces power dissipated in the primary
switch means of the converters and significantly reduces the input ripple
current and output ripple current at the input and output ports,
respectively, of a dual converter according to the present invention.
Accordingly, it is an object of the present invention to provide greater
power conversion efficiencies for the plurality of forward-converter
circuit topologies over a greater dynamic range of load current levels.
It is another object of the present invention to provide
zero-voltage-switching conditions for the primary switch means of a
forward converter topology and to provide these conditions over a large
dynamic range of load current levels.
It is another object of the present invention to minimize the input and
output ripple currents of the single-ended converter topology.
It is yet another object of the present invention to reduce the
electromagnetic interference (EMI) radiation effects of the single-ended
power converter topology.
It is still another object of the present invention to provide a low-cost,
compact converter having a low input ripple current characteristic.
It is still a further object of the present invention to provide a switch
control means for controlling the second switch means which operates in an
autonomous manner with respect to the means for controlling the primary
switch means.
These and other objects of the present invention will become apparent to
those skilled in the art from the following detailed description of the
invention and from the accompanying drawings.
BRIEF DESCRIPTION OF THE DRAWINGS
FIG. 1 is a circuit diagram of a prior art forward converter having a
series combination of a storage capacitor and auxiliary switch coupled in
parallel with the primary transformer winding according to the prior art.
FIG. 2 is a first timing diagram of key voltages and currents of the prior
art forward converter shown in FIG. 1 under light loading conditions.
FIG. 3 is a second timing diagram of key voltages and currents of the prior
art forward converter shown in FIG. 1 under heavy loading conditions.
FIG. 4 is a circuit diagram of an exemplary embodiment of a forward
converter preferably used as one of the two forward converters of the dual
active clamp converter according to the present invention.
FIG. 5 is a timing-requirement diagram showing the operation of the first
and second switch means of the exemplary embodiment of the forward
converter shown in FIG. 4.
FIG. 6 is a first timing diagram of key voltages and currents of the
exemplary embodiment of the forward converter shown in FIG. 4 under a
first switching-event sequence.
FIG. 7 is a second timing diagram of key voltages and currents of the
exemplary embodiment of the forward converter shown in FIG. 4 under a
second switching-event sequence.
FIG. 8 is a block diagram of a first embodiment of the dual active clamp
converter according to the present invention.
FIGS. 9, 9A, and 9B show more detailed partial block and schematic diagrams
of the first embodiment of the dual active clamp converter according to
the present invention.
FIG. 10 is a timing diagram of key currents and voltages of the dual active
clamp forward converter according to the present invention shown in FIG.
9.
FIGS. 11, 11A, and 11B are a more detailed embodiment of the control means
according to the present invention.
FIG. 11C is a schematic diagram of an autonomous switch control means for
the second switch means according to the present invention.
FIG. 12 is a timing diagram of key signals of the control means according
to the present invention shown in FIGS. 11A and 11B.
FIG. 13 is a second embodiment of the active-clamp forward converter
according to the present invention.
FIG. 14 is a third embodiment of the active-clamp forward converter
according to the present invention.
FIG. 15 is a set of graphs comparing the relative input ripple voltage
produced by a single active clamp forward converter and by two dual active
clamp forward converters according to the present invention.
DETAILED DESCRIPTION OF THE PRESENT INVENTION
The features of the present invention may be better appreciated and
comprehended with a more detailed description of the prior art development
of the forward converter. The basic topology of the forward converter
comprises a power transformer having a primary winding and secondary
winding, a primary switch connected in series with the primary winding,
and two commutating rectifiers which direct power from the secondary
winding to a load. A voltage source V.sub.0 supplies power to the primary
winding, and a choke inductor and a load capacitor filter the power
delivered to the load. The operation of basic forward converter is
governed by the primary switch, which causes the forward converter to
enter an ON-period when it is closed and to enter an OFF-period when it is
open. The duration of the ON-period, with respect to the duration of the
OFF-period, is varied to supply a desired amount of voltage and power to
the load.
When the primary switch is closed, magnetizing current and magnetic energy
build up in the power transformer due to the voltage applied across the
primary winding. The voltage across the primary winding is also applied to
the secondary winding by transformer action and a current in the secondary
winding is induced to flow through the commutating rectifiers towards the
load. A current related to the current in the secondary winding is
reflected back to primary winding by transformer action. The power
transformer further includes a third winding for resetting the core of the
transformer during the OFF period of the primary switch. The third winding
is coupled to the input voltage source V.sub.0 by way of a rectifier and
discharges, during the OFF-period, the magnetizing current (and energy)
built up in the power transformer during the preceding ON-period.
There are several critical operational aspects which affect the reliability
and power-conversion efficiency of the basic forward converter. One aspect
relates to the magnetic operation of the power transformer, which is
usually constructed with a core of soft ferromagnetic material of
negligible hysteresis. Due to the physical properties of the ferromagnetic
material, the core can only support a maximum amount of magnetic flux
density, B. Since the magnetic flux .PHI. is proportional to the magnetic
flux density B by the number of turns and the cross-sectional area of the
winding, the core can only support a maximum amount of magnetic flux and,
hence, magnetizing current. This amount is referred to as the saturation
flux value. Beyond the saturation flux value, the transformer ceases to
operate.
The saturation of magnetic flux limits the amount of power that the basic
forward converter may provide to the load during each ON-period. To
increase the saturation flux value, the number of turns in each winding
can be increased or the cross-sectional area can be increased. Either of
these actions, however, increases the parasitic resistances of the
windings and thereby decreases the efficiency of the basic forward
converter. As noted in U.S. Pat. No. 4,441,146 issued to Vinciarelli, the
basic forward converter only uses the first quadrant of the transformer's
B-H curve and, hence, only uses one half of the transformer's flux range.
As suggested in that patent, if both the first and third quadrant of the
transformer's B-H curve are used instead of only the first quadrant, the
transformer of a forward converter can be re-designed to reduce the
winding resistances and thereby increasing the converter's efficiency.
Another operational aspect of the basic forward converter relates to the
voltages and currents of the primary switch during switching events. The
primary switch generally comprises a semiconductor device such as a
field-effect transistor (FET) or a bipolar-junction transistor (BJT). When
switching from the OFF-period to the ON-period, it takes a finite amount
of time for the voltage across the primary switch to decrease to zero as
the current through the primary switch increases, resulting in a large
power dissipation in the primary switch. The dissipated power is equal to
the product of both the current and voltage and is zero if either quantity
is zero. The power dissipated during the switching transitions directly
reduces the conversion efficiency of the forward converter by converting
electrical energy into heat. In turn, the reliability and lifetime of the
primary switch is reduced by the heating. The present invention addresses
this power conversion loss by providing means for driving the voltage
across the primary switch means to near zero volts before the primary
switch means is closed. The condition whereby the voltage across the
primary switch means is driven to zero volts before the primary switch is
closed is herein referred to as a zero-voltage-switching condition.
While switching from the ON-period to the OFF-period, it takes a finite
amount of time for the current in the primary switch to decrease to zero
as the voltage across the primary switch means increases, resulting in a
power dissipation in the primary switch means. The power dissipation
during this transition is much less than the dissipation in the transition
from the OFF-period to the ON-period, typically between one and two orders
of magnitude less. The difference in dissipation is related to the
properties of the transistor device, either FET or BJT, and an explanation
of the difference is not necessary in order to understand the present
invention and enable one of ordinary skill in the art to make and use the
same.
Another operational aspect of the basic forward converter relates to the
voltage stress on the primary switch means during the OFF-period. During
the OFF-period, the voltage across the primary winding reverses and adds
with voltage source V.sub.0 to produce a voltage stress on the primary
switch approximately equal twice the input voltage V.sub.0, taking the
general case where the number of turns of the first and third windings are
the same. This voltage stress on the primary switch affects the
reliability of the basic forward converter directly and the conversion
efficiency indirectly. The indirect effect on efficiency is explained
below as follows. Due to the material properties of semiconductor devices,
FET and BJT devices can only withstand a predetermined maximum voltage,
known as a breakdown voltage, across their conduction terminals before the
devices are destroyed. It is well known within the semiconductor device
fabrication art that increasing the breakdown voltage of a BJT or FET
device requires increasing the "on"-resistance between the conduction
terminals of the semiconductor device. The increased "on"-resistance
lowers the conversion efficiency by increasing the power dissipation in
the primary switch during the ON-period.
To address the indirect power dissipation losses due to the
underutilization of the core and due to voltage stresses on the primary
switch means, a modification to the standard forward converter was
proposed in U.S. Pat. No. 4,441,146 issued to Vinciarelli and in an
article by B. Carsten, entitled "High Power SMPS Require Intrinsic
Reliability," PCI Proceedings, March 1982, pp. 456-471. As the present
invention builds on the teachings of prior art active clamp circuits, an
exemplary such circuit is discussed in greater detail below with reference
to FIGS. 1 through 3. Note that this modified circuit does not address the
direct power dissipation losses in the primary switch means during
transitions between the ON and OFF-periods. Such direct power dissipation
losses become more significant as the switching frequency increases since
the duration of each switching event comprises a larger fraction of each
switching cycle duration as the switching frequency increases.
The exemplary modified forward converter described above is shown at 3000
in FIG. 1. Forward converter 3000 comprises a transformer 3020 having a
primary winding 3022 and secondary winding 3024, a primary switch means
3040 (S.sub.PR) connected in series with primary winding 3022, and
rectifiers 3050 (CR1) and 3060 (CR2) which direct power from secondary
winding 3024 to a load 3074. A voltage source 3010 having a value of
V.sub.0 supplies forward converter 3000 with power and a choke inductor
3070 (L.sub.OUT) and a load capacitor 3072 (C.sub.OUT) are used to filter
the power delivered to load 3074. Additionally, converter 3000 comprises a
capacitor 3032 (C.sub.S) for representing the stray (parasitic)
capacitance associated with primary winding 3022 and primary switch means
3040. Converter 3000 further comprises a series combination of a storage
capacitor 3034 and an auxiliary switch means 3030 coupled in parallel with
primary winding 3022. The series combination is often referred to as an
active clamp because it acts as a voltage clamp which is actively (i.e.,
selectively) coupled to the transformer's windings. One terminal of the
series combination is coupled to the switched terminal of primary winding
3022 (at node V.sub.1) and the other end is coupled to a constant voltage
reference, which is the positive terminal of voltage source 3010 in this
case. Storage capacitor 3034 and auxiliary switch means 3030 comprise
means for discharging the magnetizing current in inductor 3023 during
OFF-periods when primary switch means 3040 is open. Additionally, storage
capacitor 3034 and auxiliary switch means 3030 comprise means for causing
converter 3000 to use the third quadrant of transformer 3020's B-H
characteristic. Additionally, the latter two components are used to
control the reset of the transformer core during the OFF-period, as
explained in greater detail below.
As discussed below, an imaginary inductor is used to model the effects of
the magnetizing energy of transformer 3020. The modeling inductor is
indicated in phantom as inductor 3023 in FIG. 1 and is coupled in parallel
with primary winding 3022 (and hence in parallel with the series
combination of auxiliary switch means 3030 and capacitor 3034). As such,
the magnetizing effects of transformer 3020 can be readily analyzed in the
form of a magnetizing current. The use of modeling inductor 3023 in this
manner is well known to the transformer art and is discussed in greater
detail below. Nonetheless, it is important to recognize that inductor 3023
is not a real component of converter 3000 but rather an imaginary modeling
component which aids in explaining the characteristic behavior of the
magnetizing energy in real transformer 3020.
Certain voltages and currents in converter 3000 are significant and their
symbolic notations are detailed below with reference to FIG. 1. The
voltage across primary winding 3022 is designated as V.sub.P, the voltage
across secondary winding 3024 is designated as V.sub.S. Each of these
winding voltages is referenced such that the positive terminal coincides
with the magnetic reference dot shown in FIG. 1. There are two node
voltages of significance. The node coupling the negative terminal of
primary winding 3022 to primary switch means 3040 is referred to as node 1
and the voltage at this node with respect to ground is designated as
V.sub.1. The node coupling rectifiers 3050 and 3060 to choke inductor 3070
is referred to as node 2 and the voltage at this node with respect to
ground is designated as V.sub.2.
The following currents are significant. The current flowing into primary
winding 3022 at the positive terminal is designated as I.sub.1, the
current flowing into inductor 3023 from voltage source 3010 is designated
as I.sub.M, and the total primary current flowing into both inductor 3023
and winding 3022 from voltage source 3010 is designated as I.sub.P. The
current flowing into capacitor 3032 (C.sub.S) from node V.sub.1 is
designated as I.sub.CS. Rectifiers 3050 (CR1) and 3060 (CR2) have the
current reference designations I.sub.CR1 and I.sub.CR2, respectively,
where the current references flow into the anode terminals of each
rectifier. The current flowing through choke inductor 3070 (L.sub.OUT) is
designated by the current reference designation I.sub.LOUT, where the
current reference flows toward load 3074. The current flowing through load
3074 is designated by the current reference designation I.sub.LOAD, where
the current reference flows from choke inductor 3070 to ground. The
current flowing in capacitor 3034 (C.sub.C) from switch means 3030 is
designated by the current reference designation I.sub.CC.
For the purposes of discussion, primary winding 3022 and secondary winding
3024 are taken to have the same number of turns. The dot notations for
windings 3022 and 3024 indicate their polarities. As mentioned above, an
imaginary inductor 3023 (L.sub.M) is indicated, in phantom, in the circuit
diagram for transformer 3020 to model the effects of the magnetizing flux
and magnetizing current of transformer 3020. As well known in the
transformer art, a current must be provided to one of windings of a real
transformer to provide the magnetomotive force required to overcome the
magnetic reluctance of the transformer's core. This current is known as
the magnetizing current and is generally provided to the primary winding
of the transformer. Since an ideal transformer does not have any magnetic
reluctance, magnetizing currents are not present in an ideal transformer.
In a real transformer, the magnetizing current can be electrically modeled
by a phantom inductor coupled in parallel with the primary winding of an
ideal transformer, as shown by inductor 3023. This is because the
magnetizing current is proportional to the time integral of the voltage
appearing across the primary winding of the transformer. The inductance
value L.sub.M of the phantom modeling inductor 3023 is set to a value
representative of the core reluctance of transformer 3020, as well known
in the transformer art.
The magnetizing current is a parasitic effect and does not exist in an
ideal transformer. As such, the magnetizing current component in the
primary winding circuit is not transformed into current in the secondary
winding. The current flowing in secondary winding 3024 is related to the
current flowing into primary winding 3022 and is not related to the
current flowing in inductor 3023. For the purposes of understanding the
operation of transformer 3020, primary winding 3022 and secondary winding
3024 function as an ideal transformer while inductor 3023 functions to
account for the magnetizing current in transformer 3020. As such, the
above mentioned current I.sub.P represents the current flowing into the
primary winding of a real transformer, the above mentioned current I.sub.M
represents the component of primary current I.sub.P needed to overcome the
reluctance of the real transformer, and the above mentioned current
I.sub.1 represents the component of primary current I.sub.P which is
actually transformed over to the secondary winding of the real
transformer. As a point of generality, inductor 3023 can be coupled in
parallel with secondary winding 3024 rather than with primary winding 3022
to represent the magnetizing current.
The magnetizing current causes energy to be stored in the transformer's
core when the transformer's primary winding is energized by the closing of
primary switch means 3040. This energy is stored in the form of magnetic
flux in the core and is referred to as the magnetizing flux. The
magnetizing flux must be periodically discharged to prevent the flux in
the core from reaching a saturation point, at which time the transformer
ceases to provide transformer action. The material properties of the core
limit the total amount of flux the core can store. The flux energy is
discharged by discharging the magnetizing current through one of the
transformer's windings. The discharging process is formally known as
resetting the core.
The operation of forward converter 3000 is governed by primary switch means
3040, which causes converter 3000 to enter an ON-period when switch means
3040 is closed and to enter an OFF-period when switch means 3040 is open.
Auxiliary switch means 3030 operates counter to primary switch means 3040
by opening when primary switch means 3040 closes and by closing when
primary switch means 3040 opens. The operation of converter 3000 is
explained in greater detail below with the assumption that the voltage at
the positive terminal of capacitor 3034 with respect to ground is greater
than V.sub.0, which is the voltage value of voltage source 3010.
During the ON-period, when primary switch means 3040 is closed and
auxiliary switch means 3030 is open, a voltage V.sub.0 is applied to
primary winding 3022, which is then applied to secondary winding 3024 by
transformer action. A current in secondary winding 3024 is induced to flow
through rectifier 3050 towards load 3074. Because windings 3022 and 3024
have the same number of turns, a current equal in magnitude to the current
in secondary winding 3024 is generated in primary winding 3022 by
transformer action. At the same time, a magnetizing current I.sub.M builds
up in inductor 3023 at a constant rate (i.e , the waveform of the
magnetizing current has a positive linear slope) due to the application of
a constant voltage of V.sub.0 across inductor 3023 by voltage source 3010.
Magnetizing energy builds upon in transformer 3020 due to the magnetizing
current I.sub.M.
When primary switch means 3040 is opened and auxiliary switch means 3030 is
closed to start the OFF-period, the voltage across primary winding 3022
and secondary winding 3024 abruptly reverses to a negative value since the
voltage at the positive terminal of capacitor 3034 is greater than
V.sub.0. As a result, the current flow through secondary winding 3024
abruptly stops due to the reversal in the voltage across the secondary
winding and the blocking action of rectifier 3050. In response to
secondary winding 3024, the current flow through primary winding 3022 also
abruptly stops. The magnetizing current, however, continues to flow and is
diverted from primary switch means 3040 into storage capacitor 3034 via
auxiliary switch means 3030. For the purposes of discussion and without
loss of generality, the capacitance of storage capacitor 3034 is taken to
be large enough so that the voltage across storage capacitor 3034 does not
substantially change during the OFF-period. Since a constant negative
voltage is being applied to inductor 3023 during the OFF-period, the
magnetizing current I.sub.M decreases at a constant rate, thereby acting
to reset the transformer core. If the OFF-period is sufficiently long, the
decrease in I.sub.M continues to the point where current I.sub.M reverses
direction and flows out of storage capacitor 3034 and into voltage source
3010.
The ratio of the duration of the ON-period to the time interval between
each successive ON-period is known as the duty-cycle of the converter. The
time interval between each successive ON-period is known as the switching
period T. The duty cycle is a fraction which ranges between 0.0 and 1.0,
but can be given as a percentage, which is equal to the fraction
multiplied by 100%, i.e., (ON-period)/T*100%. The duty-cycle is adjusted
to meet the power requirements of the load, with a higher duty-cycle
providing more power to the load. The duration of one ON-period and the
subsequent OFF-period is often referred to as the switching cycle. The
OFF-period is often referred to as the "fly-back" phase due to the abrupt
reversals in the voltages and changes of the currents of the transformer
windings. It is also referred to as the "reset" phase since the magnetic
energy and flux in the transformer core decreases to zero during this
time. The voltage that develops across primary switch means 3040 during
the OFF-period is greater than V.sub.0 and is referred to as the "reset"
voltage.
Under steady-state conditions, the output voltage V.sub.OUT and the
duty-cycle of the forward converter are generally related as follows. When
choke inductor 3070 continuously provides current to load 3074 throughout
the switching cycle, the output voltage V.sub.OUT is equal to the input
voltage V.sub.0 times the duty cycle, as expressed in the form of a
fraction (i.e., divided by 100%). This is because the output voltage
V.sub.OUT adjusts to a voltage level such that the time-integrated voltage
applied across choke inductor 3070 during each switching cycle is equal to
zero, under steady-state conditions. During the ON-period duration,
T.sub.ON, the voltage applied to choke inductor 3070 is (V.sub.0
-V.sub.OUT) volts. During the OFF-period duration, T.sub.OFF, the voltage
applied to choke inductor 3070 is (-V.sub.OUT) volts. Setting the
time-integrated voltage to zero (i.e., T.sub.ON (V.sub.0 -V.sub.OUT)
-T.sub.OFF V.sub.OUT =0) and solving for V.sub.OUT results in V.sub.OUT
=V.sub.0 T.sub.ON /(T.sub.ON +T.sub.OFF)=V.sub.0 *(duty-cycle/100%). In
the case where choke inductor 3070 does not continuously provide current
to load 3074, due to low load-current demand, the relationship between the
duty-cycle and the output voltage is considerably more complex and depends
upon the output current level required by load 3074. The derivation and
presentation of this more complex relationship are not necessary in order
to understand the present invention and enable one of ordinary skill in
the art to make and use the present invention. (The derivation and
presentation of the more complex relationship may be found in the forward
converter switching art.) For the purposes of presenting the present
invention and without loss of generality, the exemplary operating
conditions presented hereinbelow assume that choke 3070 continuously
provides current to load 3074 and makes use of the relationship V.sub.OUT
=(duty-cycle/100%)*V.sub.0.
To prevent the core of transformer 3020 from saturating during steady-state
operations, the time-integrated voltage across inductor 3023, and hence
across each winding of transformer 3020, should be zero for each switching
cycle, otherwise the magnetic flux in the core of transformer 3020 will
saturate within a finite number of switching cycles. In the steady-state
operation of converter 3000, the voltage V.sub.C across capacitor 3034 is
at a value which sets the time-integrated voltage across inductor 3023 to
zero during each switching cycle. This value is referred to as the
required value for V.sub.C which prevents core saturation and may be
determined as follows. During the ON-period duration T.sub.ON, a voltage
of +V.sub.0 is applied to inductor 3023 by voltage source 3010. During the
OFF-period duration T.sub.OFF, the voltage applied to inductor 3023 is the
negative of the voltage across storage capacitor 3034: -V.sub.C. In this
case, the time-integrated voltage during each switching cycle is
determined as:
Time Integrated Voltage=T.sub.ON V.sub.0 -T.sub.OFF V.sub.C(1)
Setting the time integrated voltage to zero and solving for the required
value of V.sub.C, which sets the time-integrated voltage to zero, results
in:
##EQU1##
where the symbol V.sub.CSS represents the required value of V.sub.C for
preventing core saturation during steady-state operations and the symbol
DC represents the given duty cycle.
To keep the voltage V.sub.C on storage capacitor 3034 at the required value
V.sub.CSS under the conditions of constant input voltage, constant output
voltage, and constant output load current, the flow of charge into storage
capacitor 3034 during the first half of the OFF-period must be equal to
the flow of charge out of storage capacitor 3034 during the second half of
the OFF-period so that no net accumulation or depletion of charge occurs
on storage capacitor 3034 during the OFF-period. Given that the
magnetizing current I.sub.M increases at a constant rate during the
ON-period, that I.sub.M decreases at a constant rate during the OFF-period
and that I.sub.M is continuous, the waveform for the magnetizing current
I.sub.M must be centered around a value of zero amperes in order to
prevent a net charge accumulation or depletion on storage capacitor 3034
during the OFF-period. Otherwise, the voltage V.sub.C would shift away
from the required V.sub.CSS value.
If the I.sub.M waveform is not centered around zero amperes, the voltage
V.sub.C departs from the value V.sub.CSS in order to shift the I.sub.M
waveform so that it is centered around zero amperes. For example, if the
I.sub.M waveform is initially shifted upwards and centered around a
positive current value, the shifted waveform will cause a net charge to
flow into storage capacitor C.sub.C, thereby raising the voltage V.sub.C.
In turn, the higher V.sub.C voltage causes the magnetizing current to
decrease at a faster rate during the OFF-period, thereby shifting the
I.sub.M waveform down to center the I.sub.M waveform around zero amperes
within a finite number of switching cycles. Likewise, if the I.sub.M
waveform is initially shifted downwards and centered around a negative
current value, the shifted waveform will cause a net charge to flow out of
storage capacitor C.sub.C, thereby lowering the voltage V.sub.C. In turn,
the lower V.sub.C voltage causes the magnetizing current to decrease at a
slower rate during the OFF-period, thereby shifting the I.sub.M waveform
up to center the I.sub.M waveform around zero amperes within a finite
number of switching cycles.
During power-up and during changing load conditions, the voltage value
V.sub.CSS for storage capacitor 3034 is reached by the transient charging
or discharging of storage capacitor 3034. If the voltage across storage
capacitor 3034 is less than V.sub.CSS, It takes a longer time duration to
reverse the current-flow direction of I.sub.M and, consequently, more
current flows into storage capacitor 3034 than flows out during the
OFF-period. The net current flow into capacitor 3034 during each switching
cycle raises the voltage across storage capacitor 3034 until the voltage
V.sub.CSS is reached. Likewise, if the voltage on storage capacitor 3034
is greater than V.sub.CSS, it takes a shorter time duration to reverse the
current-flow direction of I.sub.M and, consequently, more current flows
out of storage capacitor 3034 than flows in during the OFF-period. The net
current flow out of capacitor 3034 during each switching cycle lowers the
voltage across storage capacitor 3034 until the voltage V.sub.CSS is
reached. During this transient charging condition, the waveform for the
magnetizing current is not centered around a value of zero amperes.
The energy transferred out of storage capacitor 3034 during the latter part
of the OFF-period is recycled to voltage source 3010. The reversal of
current flow in the magnetizing current I.sub.M though inductor 3023
serves to reset the magnetic flux in the transformer's core at a point
inside the third quadrant of the transformer's B-H characteristic. The
utilization of the third quadrant in this manner increases the utilization
of the transformer's core and, hence, the structure of transformer 3020
may be re-designed to decrease the dissipation losses in transformer 3020,
as described above.
The steady-state waveforms for the key voltages and currents of forward
converter 3000 are shown in timing diagram 4000 in FIG. 2. For the
purposes of this discussion, each winding of transformer 3020 has the same
number of turns and has no parasitic resistance and transformer 3020 is
operated in the linear regime of its B-H characteristic. Additionally,
each rectifier has a nominal voltage drop for forward conducting current,
near zero current conduction for reverse applied voltage and has a nominal
amount of reverse-recovery current. To simplify the discussion of forward
converter 3000, the output voltage across load 3074 is taken to be
constant at a value of V.sub.0 /3. Furthermore, load 3074 is taken to be
resistive in nature without loss of generality.
The operation of primary switch means 3040 is shown in graph 4001 in FIG. 2
and the operation of auxiliary switch means 3030 is shown in graph 4009 in
FIG. 2. Graph 4001 is in the form of a time line which indicates the
ON-periods where primary switch means 3040 (S.sub.PR) is closed and the
OFF-periods where primary switch means 3040 is open. Graph 4009 is also in
the form of a time line and indicates when auxiliary switch means 3030
(S.sub.A) is closed and when auxiliary switch means 3030 is open. An
exemplary duty-cycle of 33.3%, which is consistent with an output voltage
V.sub.OUT of V.sub.0 /3, is shown for primary switch means 3040 (S.sub.PR)
in graph 4001.
The winding voltages V.sub.P and V.sub.S and the voltage V.sub.1 are shown
in graph 4002 in FIG. 2. For the primary circuit, the total primary
current I.sub.P and the current I.sub.CS into capacitor C.sub.S are shown
in graph 4003, the magnetizing current I.sub.M in inductor 3023 and the
current I.sub.1 in primary winding 3022 are shown in graph 4004, and the
current I.sub.CC flowing into storage capacitor 3034 is shown in graph
4005 in FIG. 2. For the secondary circuit, the output voltage V.sub.OUT at
load 3074 and the voltage V.sub.2 are shown in graph 4006, the currents
I.sub.CR1 and I.sub.CR2 in rectifiers 3050 (CR1) and 3060 (CR2) are shown
in graph 4007, and the current I.sub.LOUT in choke inductor 3070 and the
current I.sub.LOAD in load 3074 are shown in graph 4008 in FIG. 2. In
comparing the rectifier currents in graph 4007 against the inductor
current I.sub.LOUT in graph 4008, it may be seen that the rectifier
current I.sub.CR1 supports the inductor current I.sub.LOUT during the
ON-periods and that the rectifier current I.sub.CR2 supports the inductor
current I.sub.LOUT during the OFF-periods.
During the ON-period, the voltage across primary winding 3022 and secondary
winding 3024 is set by voltage source 3010 at a value of V.sub.0 with the
voltage V.sub.1 across primary switch means 3040 at near zero volts, as
shown in graph 4002. The application of V.sub.0 across the secondary
winding causes rectifier 3050 (CR1) to fully support the current in choke
inductor 3070. Via the transformer action of transformer 3020, the
rectifier current I.sub.CR1 is replicated in the current I.sub.1 of
primary winding 3022, as shown in graph 4004. Also during the ON-period,
the magnetizing current I.sub.M increases at a constant rate determined by
the voltage applied across inductor 3023 and the magnetizing inductance of
inductor 3023, as also shown in graph 4004. As discussed above, the
magnetizing current during steady-state conditions is centered around a
value of zero amperes, as shown in graph 4004. The total primary current
I.sub.p is equal to the sum of I.sub.1 and I.sub.M and is shown in graph
4003. During the ON-period, auxiliary switch means 3030 is open and, as
such, and the current I.sub.CC into capacitor 3034 is zero, as shown in
graph 4005.
As switch means 3040 opens to begin the OFF-period, auxiliary switch means
3030 begins to close. During this switching transition, the magnetizing
current in inductor 3023 initially flows into capacitor 3032 (C.sub.S) and
raises the voltage V.sub.1 towards the voltage on capacitor 3034. Once
auxiliary switch means 3030 closes, the magnetizing current in inductor
3023 flows into storage capacitor 3034. As mentioned above, the
capacitance value of storage capacitance 3034 is sufficiently large so
that the voltage across capacitor 3034 does not substantially change
during the duration of the OFF-period. For an exemplary duty-cycle of
33.3%, the steady-state voltage on storage capacitor 3034 is equal to
V.sub.0 /2, where V.sub.0 is the value of voltage source 3010. This value
is given by equation (2) above.
The application of -V.sub.0 /2 volts across primary winding 3022 and
secondary winding 3024 by storage capacitor 3034 causes rectifier 3060
(CR2) to fully support the current in choke inductor 3070, as shown in
graph 4007. As a result, the current in secondary winding 3024 ceases, as
shown by I.sub.CR1 in graph 4007, and the current in primary winding 3022
ceases, as shown by I.sub.1 in graph 4004. Additionally, the magnetizing
current I.sub.M in inductor 3023 decreases at a constant rate during the
OFF-period, as shown by graph 4004. The magnetizing current I.sub.M
initially flows into storage capacitor 3023. The rate of decrease in
I.sub.M is equal in magnitude to half of the rate of increase in I.sub.M
during the ON-period since the magnitude of applied voltage across
inductor 3023 during the OFF-period is half the magnitude during the
ON-period. Half way through the OFF-period, under steady-state conditions,
I.sub.M reverses sign and flows out of storage capacitor 3023. The
waveform of I.sub.M during the OFF-period is reflected in the waveforms
for the total primary current I.sub.p, as shown in graph 4003, and for the
storage capacitor current I.sub.CC, as shown in graph 4005.
Forward converter 3000 has two main advantages as compared with a basic
forward converter. First, the waveform of magnetizing current I.sub.M is
centered around zero amperes, indicating that the third quadrant of the
transformer core's B-H characteristic is being utilized. In contrast, the
magnetizing current waveform of the basic forward converter is
non-negative and is not centered around zero amperes, indicating that only
the first quadrant is used. The use of first and third quadrants by
forward converter 3000 provides a larger range of linearity in the B-H
characteristic, and hence the design of transformer 3020 can be optimized
to lower the winding resistances and reduce power dissipation, as
discussed above. Secondly, the voltage stress applied to the primary
switch means 3040 during the OFF-period is reduced compared to that across
the primary switch of a basic forward converter since the voltage applied
to primary switch means 3040 during the OFF-period is equal to 3V.sub.0
/2, assuming a 33.3% duty-cycle, rather than 2V.sub.0, as is the case for
the primary switch of a basic forward converter. Forward converter 3000,
however, does not address the power dissipation that occurs in primary
switch means 3040 between the OFF and ON-periods, as discussed above. As
can be seen in graphs 4002 and 4003 of FIG. 2, as converter 3000 switches
from the OFF-period to the ON-period, the current I.sub.p through primary
switch means 3040 is at a substantial negative value while the voltage
V.sub.1 across primary switch means 3040 is switching from 3V.sub.0 /2 to
zero volts.
To address this problem, an improvement to forward converter 3000 was
proposed in 1987 by B. Carsten, "Design Tricks, Techniques and
Tribulations at High Conversion Frequencies," HPFC, April 1987, p. 139.
The proposed improvement uses the magnetizing current present in inductor
3023 during the end of the OFF-period to reduce the voltage at node
V.sub.1 by discharging stray (parasitic) capacitor 3032. Under some
conditions, the voltage V.sub.1 can be reduced to zero volts in this
manner before primary switch means 3040 is closed, thereby significantly
reducing the power dissipation in primary switch means 3040 during
switching-on transitions. Under other conditions, however, the voltage
V.sub.1 can never be reduced to zero volts before primary switch means
3040 closes. As discussed below, the present invention identifies the
causes which prevent zero voltage switching and provides means for
overcoming these causes.
In this proposed improvement, a short delay period, or time duration, is
introduced between the end of the OFF-period and the beginning of the
ON-period where both primary switch means 3040 and auxiliary switch means
3030 remain open. This time period is shown at 4020 in FIG. 2 and is
referred to as delay period 4020 for the purposes of this discussion. As
can be seen in graph 4004, the magnetizing current I.sub.P through
inductor 3023 is negative at the beginning of delay period 4020. Since
both switch means 3040 and 3030 are open, no definite voltage is applied
across inductor 3023 and primary winding 3022. Under this condition, the
magnetizing current I.sub.M begins to flow through stray capacitor 3032,
thereby reducing the voltage V.sub.1 from 3 V.sub.0 /2 towards zero volts.
The magnetizing current continues to reduce the voltage V.sub.1 until the
voltage V.sub.1 reaches a value of V.sub.0. At this point, the voltages
V.sub.P across primary winding 3022 and V.sub.S across secondary winding
3024 become positive and the secondary circuit begins conducting current
through rectifier 3050 (I.sub.CR1). Via transformer action, a current
(I.sub.1) flows in primary winding 3022 which is equal in magnitude to the
current flowing in secondary winding 3024. This current flow in primary
winding 3022 diverts some or all of the magnetizing current away from
stray capacitor 3032. If the magnitude of the current in choke inductor
3070 is less than the magnitude of the magnetizing current I.sub.M at this
point, a portion of the magnetizing current will continue to flow into
stray capacitor 3032 and voltage V.sub.1 will continue to be decreased.
This portion of magnetizing current is equal to the magnetizing current
less the current flowing in choke inductor 3070.
As the voltage V.sub.1 is decreased, the applied voltage across primary
winding 3022 and secondary winding 3024 increases, thereby increasing the
current flow in choke inductor 3070 and secondary winding 3024. In turn,
this increase in secondary current reduces the portion of magnetizing
current which is flowing through stray capacitor 3032. At some point, the
decrease in voltage V.sub.1 ends because the current in secondary winding
3024 is increased to the point where it equals the magnetizing current in
inductor 3023. Under certain load conditions and duty-cycles, it is
possible to decrease the voltage V.sub.1 to zero volts or less, thereby
facilitating a zero-voltage-switching condition for primary switch means
3024. Such an exemplary case is shown in FIG. 2 where the voltage V.sub.1
shown in graph 4002 is decreased to a negative value during delay period
4020.
However, it is also possible that the voltage V.sub.1 cannot be reduced to
zero volts due to the loading effects on secondary winding 3024. In some
cases, the reduction of voltage V.sub.1 ends at a positive voltage. More
likely, however, the current through choke inductor 3070 is greater than
the magnetizing current I.sub.M during the beginning of delay period 4020
and oftentimes throughout delay period 4020. In this case, the magnetizing
current decreases voltage V.sub.1 only to a value of V.sub.0 volts, at
which point all of the magnetizing current I.sub.M is diverted into
secondary winding 3024, via primary winding 3022, and the voltage V.sub.1
is clamped at a value of V.sub.0. The remainder of the current flow in
choke inductor 3070 is provided by rectifier 3060, which clamps voltage
V.sub.2 near zero volts. The current conduction in both rectifiers 3050
and 3060 clamps the secondary voltage V.sub.S to zero volts and,
consequently, ensures that voltage V.sub.1 remains at a value of V.sub.0
volts. In this case, no zero-voltage switching is possible. Unfortunately,
since forward converters are designed to maximize the current delivered to
the load and to minimize the magnitude of magnetizing current, it is
rarely possible to achieve a zero-voltage-switching condition because the
load current is larger than the magnetizing current during the delay
period.
The steady-state waveforms for the key voltages and currents of forward
converter 3000 under the conditions where the current through choke
inductor 3070 is greater than the magnetizing current during the delay
period are shown in timing diagram 5000 in FIG. 3. Timing diagram 5000
comprises graphs 5001 through 5009 and a delay period 5020. Graphs 5001
through 5009 display the same information as graphs 4001 through 4009,
repectively, as shown in FIG. 2. Once voltage V.sub.1 is decreased to a
value of V.sub.0 volts at or near the beginning of delay period 5020, the
current I.sub.CR1 in secondary winding 3024 begins to flow, thereby
clamping the voltage V.sub.S on secondary winding 3024 to zero volts since
rectifier 3060 is still conducting. A voltage of zero volts is now applied
to the windings of transformer 3020, and the magnetizing current is
prevented from flowing into stray capacitor 3032. Instead, the magnetizing
current, shown in graph 5004, is diverted in primary winding 3022, as
shown by I.sub.1 in graph 5004, which sets the current in secondary
winding 3024 equal to the magnetizing current, as shown by I.sub.CR1 in
graph 5007. The remainder of the current I.sub.LOUT in choke inductor
3070, as shown in graph 5008, is conducted by rectifier 3060, as shown by
I.sub.CR2 in graph 5007. The voltage V.sub.1 is thereby clamped at a value
of V.sub.0 volts and a zero-voltage-switching condition cannot be
achieved.
The above described mechanism which diverts magnetizing current into
secondary winding 3024 and resulting voltage clamping of the transformer
windings during the delay period have thus far not been discussed nor
addressed in the prior art. The present invention has recognized this
diversion mechanism and clamping effect as a major hindrance to achieving
a zero-voltage-switching condition for primary switch means 3040 in
forward converter 3000 under all loading conditions. As discussed below,
the present invention seeks to eliminate the diverson of magnetizing
current and the resulting voltage clamping during the delay period,
thereby permitting voltage V.sub.1 to decrease to zero volts to provide a
zero-voltage-switching condition for the primary switch means.
A first embodiment of a zero-voltage-switching (ZVS) active-clamp forward
converter preferably used as one of the two forward converters of the dual
active clamp converter according to the present invention is shown at 100
in FIG. 4. A voltage source 110 (V.sub.0) provides electrical energy to
forward converter 100. Forward converter 100 comprises a transformer 120
(T1) having a primary winding 122 and secondary winding 124, and a primary
switch means 140 (S.sub.PR) coupled in series with primary winding 122 and
voltage source 110 for selectively coupling energy from voltage source 110
to transformer 120. Forward converter 100 further comprises a storage
capacitor 134 (C.sub.C) and a first switch means 130 (S.sub.1) for
selectively controlling the current flow through storage capacitor 134.
First switch means 130 and storage capacitor 134 are coupled in series and
the resulting series combination is coupled in parallel with either the
primary winding of transformer 120 (as shown in FIG. 4) or the secondary
winding of transformer 120 (as shown in FIG. 13). The series combination
is often referred to as an active clamp because it acts as a voltage clamp
which is actively (i.e., selectively) coupled to the transformer's
windings. Additionally, converter 100 includes a capacitor 132 (C.sub.S)
coupled between the switched terminal of primary winding 122 and ground
for representing the combined stray (parasitic) capacitances of primary
winding 122, primary switch means 140, and first switch means 130. It may
be appreciated that each of primary winding 122, primary switch means 140,
and first switch means 130 may include a parasitic capacitance, resulting
from the non-ideal elements used in implementing each of these components.
The parasitic capacitance of each of these components may, for example, be
represented by a parasitic capacitor for each component terminal, each
parasitic capacitor being coupled between its respective component
terminal and ground. As these capacitances are parasitic in nature, their
exact values are not precisely known and may, in fact, vary considerably
in manufacturing. In a further embodiment of converter 100, capacitor 132
further comprises a fixed-value capacitor in addition to the parasitic
capacitances so as to provide a more predictable capacitance value for
capacitor 132, thereby increasing the manufacturing consistency of
converter 100.
As discussed above with reference to converter 3000, an imaginary inductor
is used to model the effects of the magnetizing energy of transformer 120.
The modeling inductor is indicated in phantom as inductor 123 in FIG. 4
and is coupled in parallel with primary winding 122. As such, the
magnetizing effects of transformer 120 can be readily analyzed in the form
of a magnetizing current. The use of modeling inductor 123 in this manner
is well known to the transformer art and not a real component of converter
100, but rather an imaginary modeling component which aids in explaining
the characteristic behavior of the magnetizing energy in real transformer
120.
For directing and controlling the flow of power from transformer 120 to an
output load 174, forward converter 100 further comprises a second switch
means 150 coupled in series with the secondary winding for controlling the
flow of current to load 174, and a rectifier 160 (CR2) coupled in parallel
with the series combination of second switch means 150 and secondary
winding 124. Converter 100 further includes a choke inductor 170
(L.sub.OUT) coupled between second switch means 150 and load 174, and a
load capacitor 172 (C.sub.OUT) coupled in parallel with load 174. The
current through load 174 is represented by the symbol I.sub.LOAD and, for
the purposes of discussion and without loss of generality, is taken to be
constant. Choke inductor 170 (L.sub.OUT) and load capacitor 172
(C.sub.OUT) provide means for filtering the power delivered to load 174.
Forward converter 100 further comprises a switch control means 180 for
generating signals to control the operation of primary switch means 140,
first switch means 130, and second switch means 150. Switch control means
180 comprises a port 181 coupled to primary switch means 140, a port 182
coupled to first switch means 130, and a port 183 coupled to second switch
means 150. Port 181 transmits a signal which controls the operation of
primary switch means 140, port 182 transmits a signal which controls the
operation of first switch means 130, and port 183 transmits a signal which
controls the operation of second switch means 150.
Certain voltages and currents in converter 100 are significant and their
symbolic notations are detailed below with reference to FIG. 4. The
voltage across primary winding 122 is designated as V.sub.p, and the
voltage across secondary winding 124 is designated as V.sub.S. Each of
these winding voltages is referenced such that the positive terminal
coincides with the magnetic reference dot shown in FIG. 4. There are two
node voltages of significance. At the node coupling the negative terminal
of primary winding 122 to primary switch means 140, the voltage at this
node with respect to the negative terminal of voltage source 110 is
designated as V.sub.1 and the node itself is referred to as node V.sub.1.
At the node coupling second switch means 150 and rectifier 160 to choke
inductor 170, the voltage at this node with respect to the negative
terminal of secondary winding 124 is designated as V.sub.2 and the node
itself is referred to as node V.sub.2.
The following currents in converter 100 are significant. The current
flowing into primary winding 122 at the positive terminal is designated as
I.sub.1, the current flowing into inductor 123 from voltage source 110 is
designated as I.sub.M, and the total primary current flowing into both
inductor 123 and winding 122 from voltage source 110 is designated as
I.sub.p. The current flowing into stray capacitor 132 (C.sub.S) from node
V.sub.1 is designated as I.sub.CS and the current flowing into storage
capacitor 134 (C.sub.C) from node V.sub.1 is designated as I.sub.CC. The
current flowing into second switch means 150 from secondary winding 124 is
designated as I.sub.S2 and the current flowing out of the cathode terminal
of rectifier 160 is designated as I.sub.CR2. The current flowing through
choke inductor 170 (L.sub.OUT) is designated by the current reference
designation J.sub.LOUT, where the current reference flows toward load 174.
The current flowing through load 174 is designated by the current
reference designation I.sub.LOAD, where the current reference flows from
choke inductor 170 to ground.
For the purposes of discussion and without loss of generality, primary
winding 122 and secondary winding 124 are taken to have the same number of
turns. The dot notations for windings 122 and 124 indicate their
polarities. An inductor 123 (L.sub.M) is included, in phantom, in the
circuit diagram for transformer 120 to model the effects of the
magnetizing flux and magnetizing current of transformer 120, as discussed
above. As mentioned above, a magnetizing current must be provided to one
of windings of a real transformer to provide the magnetomotive force
required to overcome the magnetic reluctance of the transformer's core.
For a real transformer, the magnetizing current can be electrically
modeled by a phantom inductor coupled in parallel with the primary winding
of an ideal transformer, as shown by inductor 123. As such, the above
mentioned current I.sub.p represents the current flowing into the primary
winding of a real transformer, such as transformer 120, the above
mentioned current I.sub.M represents the component of primary current
I.sub.p needed to overcome the magnetic reluctance of the real
transformer, and the above mentioned current I.sub.1 represents the
component of primary current I.sub.p which is actually transformed over to
the secondary winding of the real transformer.
As with forward converter 3000, the operation of forward converter 100 is
governed by primary switch means 140, which causes converter 100 to enter
an ON-period when switch means 140 is closed and to enter an OFF-period
when switch means 140 is open. First switch means 130 operates
substantially counter to primary switch means 140 by closing substantially
when primary switch means 140 opens to start an OFF-period and by opening
prior to when primary switch means 140 closes to start an ON-period.
Second switch means 150 operates substantially synchronously with primary
switch means 140 by closing substantially when primary switch means 140
closes and by opening substantially when primary switch means 140 opens.
The operation of converter 100 is explained in greater detail below with
the assumption that the voltage at the positive terminal of capacitor 134
with respect to ground is greater than V.sub.0, which is the voltage value
of voltage source 110.
The operations of first switch means 130 and second switch means 150 are
discussed in greater detail with reference to a timing diagram shown at
200 in FIG. 5, which shows the timing requirements for the operations of
first and second switch means 130,150 with respect to primary switch means
140. A single switching cycle having an ON-period followed by an
OFF-period is shown in graph 201. The opening and closing operations of
primary switch means 140, which define the ON- and OFF-periods, is shown
in graph 205 in FIG. 5. The high sections of graph 205 indicate the time
durations where primary switch means 140 is closed and the low sections
indicate the time durations where primary switch means 140 is open. The
operation and timing requirements for first switch means 130 is shown in
graph 210 and the operation and timing requirements for second switch
means 150 is shown in graph 220 in FIG. 5.
Shown in graph 210 is a timing window 212 in which switch control means 180
closes first switch means 130. Timing window 212, as well as all other
timing windows shown in timing diagram 200, is illustrated using a series
of parallel skew lines (/). In the preferred embodiment of converter 100,
timing window 212 is of a predetermined length and immediately follows the
end of the ON-period. Timing window 212 does not preferably intersect with
the ON-period. This is because, as discussed in greater detail below, the
delay in closing first switch means 130 with respect to the opening of
primary switch means 140 may be used to create a zero-voltage-switching
condition for first switch means 130.
Also shown in graph 210 is a predetermined time delay 214 in which first
switch means 130 is opened before primary switch means 140 is closed. As
discussed below in greater detail, the time delay in closing first switch
means 130 with respect to the opening of primary switch means 140 in
timing window 212 may be used to provide a zero-voltage-switching
condition for first switch means 130 and the time delay 214 between the
opening of first switch means 130 and the closing of primary switch means
140 provides a zero-voltage-switching condition for primary switch means
140. As shown in graph 220, second switch means 150 is open during time
delay 214 and, thereby, prevents the loading on secondary winding 124 from
diverting the magnetizing current away from stray capacitor 132.
Shown in graph 220 is a timing window 222 in which switch control means 180
closes second switch means 150. In the preferred embodiment of converter
100, timing window 222 is of a predetermined length and immediately
follows the end of the OFF-period. Timing window 222 preferably does not
intersect with the OFF-period. This is because, as discussed in greater
detail below, the delay in closing second switch means 150 with respect to
the closing of primary switch means 140 may be used to allow primary
switch means 140 to reach a full conduction state before primary switch
means 140 has to conduct the reflected secondary current, thereby
preventing an excessive secondary load current from pulling primary switch
means 140 out of a zero-voltage switching condition.
Also shown in graph 220 is a timing window 224 in which switch control
means 180 opens second switch means 150. In the preferred embodiment of
converter 100, timing window 224 is of a predetermined length which starts
during the ON-period and ends before the closing of first switch means 130
in timing window 212. Typically, second switch means 150 is opened
synchronously with the opening of primary switch means 140. However, as
discussed in greater detail below, the opening of second switch means 150
after the opening of primary switch means 140 may be used to direct the
current in secondary winding 124, as reflected in primary winding 122,
into stray capacitor 132 before first switch means 130 is closed. This
reduces the amount of magnetizing current required to charge capacitor 132
and is useful at low duty-cycle conditions where the magnitude of the
magnetizing current is small. As a result of conserving the magnetizing
current at low duty-cycles, the range of duty-cycles over which a
zero-voltage-switching condition for primary switch means 140 can occur is
extended.
In the alternative, second switch means 150 may be opened before the
opening of primary switch means 140, as shown at the dotted line 226 in
graph 220. This opening may be used to reduce the current in secondary
winding 124 and the corresponding reflected current in primary winding 122
before primary switch means 140 is opened. In such a case, only the
magnetizing current component remains in primary winding 122 when primary
switch means 140 is opened, thereby reducing the power dissipation and
current stresses on primary switch means 140. The reduction of such
stresses is important during high-load (i.e., high duty-cycle) conditions.
Therefore, the opening of second switch means 150 in timing window 224 may
be varied to minimize the power dissipation losses in primary switch means
140.
Also shown in timing diagram 200 is a first switching-event sequence 230
and a second switching-event sequence 240, which are used later with
reference to FIGS. 6 and 7 in discussing the operation of converter 100.
Sequences 230 and 240 each divides a single switching cycle into six
continuous, non-overlapping segments: segments 1, 2, 3, 4, 5, and 6 for
sequence 230 and segments 1, 2', 3', 4', 5 and 6 for sequence 240. The
boundary of each segment is defined by a switching event in one of the
switch means, 130, 140, and 150. Sequence 230 corresponds to the case
where second switch means 150 is opened after primary switch means 140 is
opened, as shown in window 224, and sequence 240 corresponds to the case
where second switch means 150 is opened before primary switch means 140 is
opened, as indicated by line 226.
Exemplary qualitative steady-state waveforms for the key voltages and
currents of forward converter 100 under switching-event sequences 230 and
240 are shown in a timing diagram 300 in FIG. 6 and a timing diagram 400
in FIG. 7, respectively. For the purposes of this discussion and without
loss of generality, it is assumed that each winding of transformer 120 has
the same number of turns and has no parasitic resistance and transformer
120 is operated in the linear regime of its B-H characteristic.
Additionally, rectifier 160 has a nominal voltage drop for forward
conducting current, a near zero current conduction for reverse applied
voltage and a nominal amount of reverse-recovery current. To simplify the
discussion of forward converter 100 and without loss of generality, the
output voltage across load 174 is taken to be constant at a value of
approximately V.sub.0 /3 and the voltage on storage capacitor 134 is taken
to be substantially constant at a value of approximately V.sub.0 /2.
Without loss of generality, load 174 is taken to be resistive in nature
and an exemplary duty-cycle of 33.3%, which is consistent with an output
voltage V.sub.OUT of V.sub. 0 /3, is used for primary switch means 140.
In FIG. 6, the operation of primary switch means 140 is shown in graph 301,
the operation of first switch means 130 is shown in graph 302, and the
operation of second switch means 150 is shown in graph 303. Graphs 301-303
are each in the form of a time line waveform which indicates that its
corresponding switch means is closed when the waveform is high and that
its corresponding switch means is open when the waveform is low. Sequence
230 of FIG. 5 is reproduced as sequence 304 in FIG. 6. Likewise, in FIG.
7, the operation of primary switch means 140 is shown in graph 401, the
operation of first switch means 130 is shown in graph 402, and the
operation of second switch means 150 is shown in graph 403. Graphs 401-403
are each in the form of a time line waveform which indicates that its
corresponding switch means is closed when the waveform is high and its
corresponding switch means is open when the waveform is low. Sequence 240
of FIG. 5 is reproduced as sequence 404 in FIG. 7. In order to more easily
show the transitions of key voltages and current during segments 1, 3/3',
4/4' and 6, the time-scale during each of these segments is expanded by
roughly an order of magnitude with respect to the time-scale used during
segments 2/2', and 5 (i.e., the waveforms are "stretched out" during
segments 1, 3/3', 4/4' and 6).
In FIG. 6, the winding voltages V.sub.P and V.sub.S and the voltage V.sub.1
are shown in graph 310. For the primary circuit, the total primary current
I.sub.P and the current I.sub.CS into capacitor C.sub.S are shown in graph
320, the magnetizing current I.sub.M in inductor 123 and the current
I.sub.1 in primary winding 122 are shown in graph 330, and the current
I.sub.CC flowing into storage capacitor 134 is shown in graph 340 in FIG.
6. For the secondary circuit, the output voltage V.sub.OUT at load 174 and
the voltage V.sub.2 are shown in graph 350, the currents I.sub.S2 and
I.sub.CR2 in second switch means 150 and rectifier 160 (CR2) are shown in
graph 360, and the current I.sub.LOUT in choke inductor 170 and the
current I.sub.LOAD in load 174 are shown in graph 370 in FIG. 6.
Likewise in FIG. 7, the winding voltages V.sub.P and V.sub.S and the
voltage V.sub.1 are shown in graph 410. For the primary circuit, the total
primary current I.sub.P and the current I.sub.CS into capacitor C.sub.S
are shown in graph 420, the magnetizing current I.sub.M in inductor 123
and the current I.sub.1 in primary winding 122 are shown in graph 430, and
the current I.sub.CC flowing into storage capacitor 134 is shown in graph
440 in FIG. 7. For the secondary circuit, the output voltage V.sub.OUT at
load 174 and the voltage V.sub.2 are shown in graph 450, the currents
I.sub.S2 and I.sub.CR2 in second switch means 150 and rectifier 160 (CR2)
are shown in graph 460, and the current I.sub.LOUT in choke inductor 170
and the current I.sub.LOAD in load 174 are shown in graph 470 in FIG. 7.
During segment 1 of sequences 230 (cf., FIG. 6) and 240 (cf., FIG. 7),
primary switch means 140 is closed and both first switch means 130 and
second switch means 150 are open. As discussed below, the voltage across
primary switch means 140 has been driven to substantially zero volts in
the preceding segment, segment 6, and a zero-voltage-switching condition
exits across primary switch means 140 when it closes. The
zero-voltage-switching condition is shown at 311 and 313 of graph 310 in
FIG. 6 and at 411 and 413 of graph 410 in FIG. 7. During segment 1, a
voltage of V.sub.0 appears across primary winding 122, secondary winding
124, and magnetizing inductor 123. In this segment, only the magnetizing
current I.sub.M flows in inductor 123 and primary switch means 140, since
second switch means 150 is open. The magnetizing current increases at a
constant rate and, as in converter 3000, increases from a negative value.
In a preferred embodiment of converter 100, primary switch means 140
comprises a transistor switching device and the duration of segment 1 is
sufficiently long to allow the transistor switching device to reach a full
conducting state (i.e., lowest conducting resistance) after being switched
on at the beginning of segment 1. The conduction of current in secondary
124 is held off until segment 2/2', at which time the transistor switching
device of primary switch means 140 has lowered its ON-resistance to the
point that the addition of the reflected secondary current will not
substantially raise the voltage across the device's conducting terminals
and, hence, will not increase the power dissipation in the device.
During segment 2 of sequence 230 (cf., FIG. 6) and segment 2' of sequence
240 (cf., FIG. 7), second switch means 150 is closed. During segments 2
and 2', the voltage of V.sub.0 appearing across secondary winding 124 and
the closing of switch means 150 causes a current to flow through secondary
winding 124 and a corresponding current to flow in primary winding 122,
due to transformer action. The magnetizing current in inductor 123
continues to increase at a constant rate due to the application of a
constant voltage of V.sub.0 across inductor 123 by voltage source 110.
During this segment, power is transferred to the secondary circuit from
the primary circuit.
In sequence 230 (cf., FIG. 6), the power transfer continues until primary
switch means 140 is opened, when segment 3 begins. During segment 3, the
magnetizing current I.sub.M flows into stray capacitor 132, shown in FIG.
4. This flow raises the voltage V.sub.1 and, at the same time, lowers the
voltage across primary winding 122 and secondary winding 124. The current
in secondary winding 124 is governed by choke inductor 170 and continues
to flow because the voltage across secondary winding 124, V.sub.S,
continues to be positive. The positive value for V.sub.S prevents
rectifier 160 from conducting the current in choke inductor 170. As a
result, the secondary current continues to be reflected in primary winding
122, which adds constructively with the magnetizing current I.sub.M to
charge stray capacitor 132.
The voltage across stray capacitor 132 continues to rise during segment 3
(cf., FIG. 6) until it reaches a value of V.sub.0 volts, at which time
second switch means 150 is opened to start segment 4. If second switch
means 150 is not opened when the voltage V.sub.1 across stray capacitor
132 reaches V.sub.0 volts, the voltage V.sub.S across secondary winding
124 would be clamped to substantially zero volts by rectifier 160. As a
result, the voltage V.sub.1 would be clamped to V.sub.0 volts and the
magnetizing current would be diverted to secondary winding 124. In segment
4, the magnetizing current I.sub.M continues to charge stray capacitor 132
until the voltage V.sub.1 substantially reaches the voltage on the
positive terminal of storage capacitor 134 at the end of segment 4. First
switch means 130 may then be closed with zero volts across its conduction
terminals, a zero-voltage-switching condition, to start segment 5. In FIG.
6, the zero-voltage-switching condition for switch means 130 is shown at
312 in graph 310, where voltage V.sub.1 reaches the voltage on the
positive terminal of storage capacitor 134. For an exemplary duty-cycle of
33.3%, the voltage on the positive terminal of storage capacitor 134 is
equal to 3/2 V.sub.0.
For sequence 240 (cf., FIG. 7), the power transfer continues during segment
2' until second switch means 150 is opened. The opening ends segment 2'
and begins segment 3'. During segment 3', the reflection of secondary
current in primary winding 122 ends. The magnetizing current I.sub.M,
however, continues to flow and increases in value due to the continuing
application of V.sub.0 volts across magnetizing inductance 123.
Nonetheless, the current in primary switch means 140 is significantly
reduced so that primary switch means 140 may be opened with lower current
stresses and lower power dissipation. This opening of primary switch means
140 ends segment 3' and begins segment 4'.
During segment 4' (cf., FIG. 7), the magnetizing current I.sub.M flows into
stray capacitor 132, which raises the voltage V.sub.1. Segment 4' ends
when the voltage V.sub.1 substantially reaches the voltage on the positive
terminal of storage capacitor 134. At the end of segment 4', first switch
means 130 may be closed with zero volts across its conduction terminals, a
zero-voltage-switching condition, to start segment 5. In FIG. 7, the
zero-voltage-switching condition is shown at 412 in graph 410, where
voltage V.sub.1 reaches the voltage on the positive terminal of storage
capacitor 134. For an exemplary duty-cycle of 33.3%, the voltage on the
positive terminal of storage capacitor 134 is equal to 3/2 V.sub.0.
During segment 5 for both sequences 230 (cf., FIG. 6) and 240 (cf., FIG.
7), primary switch means 140 and second switch means 150 are opened and
first switch means 130 is closed. The voltage across primary winding 122
and secondary winding 124 is set at a negative value equal to the voltage
V.sub.C across storage capacitor 134, approximately -V.sub.0 /2. The
magnetizing current I.sub.M is diverted away from charging stray capacitor
132 and flows into storage capacitor 134. In the preferred embodiment of
converter 100 and without loss of generally, the capacitance of storage
capacitor 134 is large enough so that the voltage across storage capacitor
134 does not substantially change during segment 5. (The zero-voltage
switching on primary switch means 140 and the other benefits of the
present invention could be maintained with smaller capacitance values for
storage capacitor 134, even though this would cause voltage variations
across capacitor 134 during segment 5.) Since a constant negative voltage
is being applied to inductor 123 during segment 5, the magnetizing current
I.sub.M decreases at a constant rate, thereby acting to reset the
transformer core. The duration of segment 5 is sufficiently long that the
magnetizing current I.sub.M reverses direction and flows out of storage
capacitor 134 into voltage source 110. The magnitude of the current flow
I.sub.M out of storage capacitor 134 continues to grow until first switch
means 130 opens, at which time segment 6 begins.
In segment 6, the magnetizing current is diverted away from storage
capacitor 134 towards stray capacitor 132, where it discharges stray
capacitor 132. As a result, the voltage V.sub.1 is reduced in value
towards zero volts. Unlike converter 3000, the secondary circuit in
converter 100 is prevented from interfering with the discharging of stray
capacitor 132 since second switch means 150 is opened during segment 6. At
the end of segment 6, voltage V.sub.1 is decreased to a value of zero
volts and primary switch means 140 is closed with a zero-voltage-switching
condition to start segment 1. Given the small capacitance value of stray
capacitor 132 and the typically large magnitude value of the magnetizing
current I.sub.M during segment 6, the magnitude of I.sub.M only decreases
slightly during segment 6.
As in forward converter 3000, the time-integrated voltage across inductor
123 should be zero for each switching cycle during the steady-state
operation of converter 100 to prevent the core of transformer 120 from
saturating. In the steady-state operation of converter 100, the voltage
V.sub.C across capacitor 134 is at a value which sets the time-integrated
voltage across inductor 123 to zero during each switching cycle. This
value is referred to as the required value for V.sub.C which prevents core
saturation and is similar to the required value V.sub.CSS determined for
forward converter 3000. Given the additional segments present for
converter 100, specifically segments 1, 3/3', 4/4' and 6, the calculation
of the exact value for V.sub.CSS is more complex. In practice, however,
the duration of segments 1, 3/3', 4/4' and 6 are extremely small compared
to the duration of segments 2/2' and 5, and the above equation (2) for
V.sub.CSS in converter 3000 serves as a good approximation for V.sub.CSS
in converter 100.
To keep the voltage V.sub.C on storage capacitor 134 at the required value
V.sub.CSS under the conditions of constant input voltage, constant output
voltage, and constant output load current, the flow of charge into storage
capacitor 134 during the first half of segment 5 must be equal to the flow
of charge out of storage capacitor 134 during the second half of segment 5
so that no net accumulation or depletion of charge occurs on storage
capacitor 134 during the OFF-period. As discussed above, this same
requirement was found for converter 3000. As in converter 3000, the
waveform for the magnetizing current I.sub.M in converter 100 under
steady-state conditions is centered around a value of zero amperes in
order to prevent a net charge accumulation or depletion on storage
capacitor 134 during the OFF-period. Otherwise, the voltage V.sub.C would
shift away from the required V.sub.CSS value.
As found in converter 3000, the voltage V.sub.C and the magnetizing current
I.sub.M interact so as to center the I.sub.M waveform around a value of
substantially zero amperes. Additionally, the voltage value V.sub.CSS for
storage capacitor 134 is reached during power-up conditions by transient
charging or discharging of storage capacitor 134 in the same manner found
in converter 3000. Also, the energy transferred out of storage capacitor
134 during the latter part of segment 5 is recycled to voltage source 110.
The reversal of current flow in the magnetizing current I.sub.M though
inductor 123 serves to reset the magnetic flux in the transformer's core
at a point inside the third quadrant of the transformer's B-H
characteristic.
The following exemplary component parameters and operating conditions of
converter 100 are provided as an aid in practicing the present invention.
In an exemplary embodiment of converter 100, the input voltage (V.sub.0)
is set at 300 V to deliver a typical output current of 1.0 A at an output
voltage (V.sub.OUT) of approximately 100 V. A switching frequency of 400
kHz is used in this exemplary embodiment with a typical duty-cycle of
approximately 45%. Primary switch means 140 comprises a field-effect
transistor having a breakdown voltage of 800 V and an on-resistance of 3
ohms (part number IRFBE30 from International Rectifier Corporation), first
switch means 130 comprises a field-effect transistor having a breakdown
voltage of 800 V and an on-resistance of 6 ohms (part number IRFBE20 from
International Rectifier Corporation), and second switch means 150
comprises a field-effect transistor having a breakdown of 600 V and an
on-resistance of 2.2 ohms (part number IRFBC30 from International
Rectifier Corporation). In this exemplary embodiment, storage capacitor
134 (C.sub.C ) has a capacitance of approximately 2200 pF, stray capacitor
132 (C.sub.S) has a capacitance of approximately 140 pF, and transformer
120 has an effective magnetizing inductance (L.sub.M of inductor 123) of
approximately 600 .mu.H. With this value of magnetizing inductance and
under the above operating conditions, the magnetizing current has a
peak-to-peak value of approximately 0.6 A.
It may now be appreciated that the following modifications of forward
converter 100 shown in FIG. 4 are equivalent to the specific topology
shown in FIG. 4. First, it may be appreciated that the series combination
of storage capacitor 134 and first switch mean 130 may be coupled in
parallel with secondary winding 124 instead of primary winding 122. This
is shown and discussed in greater detail below with reference to FIG. 13.
In the present invention, the coupling of the series combination must be
placed before second switch means 150. Second, it may be appreciated that
the negative terminal of storage capacitor 134 may be coupled to any
constant voltage reference, such as ground. In the case where the negative
terminal is coupled to ground, the current flow through capacitor 134
returns back to transformer 120 by way of voltage source 110. In this
sense, it may be said that the series combination is coupled in parallel
with the primary winding. Third, it may be appreciated that the positions
of storage capacitor 134 and first switch means 130 in the series
combination may be reversed. Fourth, it may be appreciated that rectifier
160, choke inductor 170, and output capacitor 172 are not essential in
providing zero-voltage-switching conditions for primary switch means 140
and for first switch means 130.
It may be appreciated that switch control means 180 shown in FIG. 4 may
comprise a digital microprocessor for generating the control signals for
primary switch means 140, first switch means 130, and second switch means
150 according to the timing constraints detailed in timing diagram 200
shown in FIG. 5. Given the inductance L.sub.M of the magnetizing inductor
123 and capacitance C.sub.S of stray capacitor 132, the duration of the
segments 1, 2/2', 3/3', 4/4', 5 and 6 required for providing zero-voltage
switching can be calculated as a function of duty-cycle for the
steady-state operation of converter 100. These values may be stored in a
conventional memory accessible to the microprocessor and the
microprocessor may select the appropriate value for each segment as
dictated by the then current value of the duty-cycle. In a further
microprocessor-based embodiment of switch control means 180, it may be
appreciated that active voltage sensors may be coupled to primary switch
means 140 and first switch means 130 for detecting zero-voltage conditions
across their respective switch means. The microprocessor of switch control
means 180 may then use this information in setting the duration of
segments 3/3', 4/4' and 6 rather than using predetermined stored values
for such segments.
As a less complex approach, which is more fully discussed in U.S. patent
application Ser. No. 07/668,587, each of switch means 130 and 140
comprises a switch transistor coupled in parallel with a rectifier (e.g.,
the body diode of a FET transistor). Each switching transistor is
responsive to the control signal provided to its corresponding switch
means 130 and 140. Each rectifier is oriented to enable current to be
conducted through its respective switch means in the desired direction
when a zero-voltage switching condition occurs across its corresponding
switch means, which often occurs before the corresponding switching
transistor is rendered conductive by its control signal. As such, each
rectifier provides means for detecting a zero-voltage-switching condition
across its switch means and means for initially conducting current through
its switch means upon the occurrence of a zero-voltage-switching condition
across the switch means. This allows the timing constraints for generating
the control signals to the switching transistors to be relaxed, which
enables switch control means 180 to be implemented with circuitry which is
substantially less complex than the above-described microprocessor-based
embodiment. In this approach, the control signal to the second switch
means is also generated in a less complex manner, being a delayed version
of the control signal provided to the switching transistor of primary
switch means 140. An example of such an approach is described in greater
detail below with reference to FIGS. 9-14.
Having thus described an exemplary embodiment of a zero-voltage-switching
forward converter preferably used as one of the two forward converters of
the dual active clamp (DAC) forward converter according to the present
invention, the DAC forward converter is now described. The DAC forward
converter according to the present invention comprises two ZVS forward
converters 100 coupled in parallel and operated in a manner so as to
substantially reduce the input ripple current of the DAC converter. This
enables the resultant compact size and low cost of the active clamp
forward converter to be exploited for low-noise and high efficiency power
supply applications. In one embodiment of the present invention, two ZVS
active-clamp forward converters are operated out of phase by substantially
180 degrees, thereby reducing the input and output ripple currents.
A block diagram of a first embodiment of the dual active clamp forward
converter according to the present invention is shown at 800 in FIG. 8.
Dual active clamp (DAC) converter 800 includes an input port 801 for
receiving a source of input power to be converted and an output port 802
for providing converter power derived from input port 801. Additionally,
DAC converter 800 comprises an input capacitor 803 coupled across input
port 801 and an output capacitor 172 coupled across output port 802. DAC
converter 800 further comprises a first forward converter 100, as
described above with reference to FIG. 4, a second forward converter 100',
and a control means 820 for coordinating the operation of converters 100
and 100'. In order to more clearly show the arrangement of converters 100
and 100', terminals and ports have been included for each of converters
100 and 100', as described below. In a preferred embodiment of the present
invention, the components and structure of second converter 100' are
essentially identical to the components and structure of first converter
100. DAC converter 800 further includes a first control bus 805 for
coupling switch control signals from switch control means 820 to first
converter 100, and a second control bus 806 for coupling switch control
signals from switch control means 820 to second converter 100'. Each
control bus 805 and 806 includes a control signal for the primary switch
means, first switch means, and second switch means of converter 100 and
100', respectively.
Converters 100 and 100' include input terminals 101 and 101' coupled to the
positive terminal of input port 801 and input terminals 102 and 102'
coupled to the negative terminal of port 802. Terminals 101 and 102
comprise an input port for receiving an input source of power for
converter 100 from input port 801, and terminals 101' and 102' comprise an
input port for receiving an input source of power for converter 100' from
input port 801. Converters 100 and 100' also include output terminals 107
and 107' coupled to the positive terminal of output port 802 and output
terminals 108 and 108' coupled to the negative terminal of port 802.
Terminals 107 and 108 comprise an output port for providing power to a
load output port 802, and terminals 107' and 108' comprise an output port
for providing power to port 802. As such, the input and output power paths
of converters 100 and 100' are coupled in parallel. Additionally, each of
converters 100 and 100' includes a port 106 and 106', respectively, for
providing to control means 820 a signal representative of the
instantaneous input current drawn by the converter. These signals are used
by control means 820 in generating the control signals for the switch
means of converters 100 and 100'.
FIG. 9 shows a more detailed partial block and schematic diagram of the
first embodiment of DAC converter 800 where the structure of converter 100
and 100' and their coupling to control means 820 is more clearly shown. As
indicated above, converter 100' preferably has a structure which is
essentially the same as that of converter 100. Accordingly, the reference
numbers for the components of second converter 100' are the same as the
corresponding reference numbers of converter 100 with the exception that a
prime indication "'" has been appended to each of the reference numbers of
second converter 100'.
First converter 100 comprises transformer 120, including windings 122 and
124 and magnetizing inductor 123, storage capacitor 134, primary switch
means 140, first switch means 130, second switch means 150, rectifier 160,
inductor 170, and stray capacitor 132, all as described above. These
components of first converter 100 are coupled to one another as described
above with reference to FIG. 4. Additionally, the output terminal of
filter inductor 170 is coupled to output terminal 107 and the negative
terminal of secondary winding 124 (i.e., the secondary-side ground) is
coupled to output terminal 108. First converter 100 further includes an
input port 103 for receiving a control signal from control means 820 for
controlling primary switch means 140, an input port 104 for receiving a
control signal from control means 820 for controlling first switch means
130, and an input port 105 for receiving a control signal from control
means 820 for controlling second switch means 150. These control signals
are provided by way of bus 805 (bus 806 for converter 100'). Not shown in
FIG. 9 are a voltage source 110 and a load 174 for converters 100 and
100'. However, it may be appreciated that a voltage source 110 may be
coupled to input port 801 and a load 174 may be coupled to output port
802.
First converter 100 further comprises a current sense transformer 135 for
sensing the current flowing into the primary circuit of first converter
100. Sense transformer 135 includes a primary winding 136 having a first
terminal coupled to terminal 101 and a second terminal coupled to primary
winding 122 of power transformer 120. Sense transformer 135 further
includes a secondary winding 137 having a first terminal coupled to ground
and a second terminal coupled to a current sense output port 106 of first
converter 100. Sense transformer 135 provides a current at port 106 which
is proportional to the current being drawn from the input supply at port
801 by converter 100. As described below in greater detail, the current
signal at port 106 is used by control means 820 in controlling the time
duration switch means 140 and 130 are closed. In a preferred embodiment of
the present invention, sense transformer 135 has a turns ratio of 1:40
(primary:secondary) and the current provided to port 106 is substantially
1/40th of the current being drawn by the primary circuit of converter 100.
As such, very little voltage is dropped across sense transformer 135,
typically less than 25 mV. Thus, it may be assumed that the voltage drop
across current sense transformer 135 does not appreciable affect the
operation of first converter 100, as described above.
In the first converter 100 shown in FIG. 9, primary switch means 140
comprises a parallel combination of a field-effect transistor 543 and
rectifier 541, and first switch means 130 comprises a parallel combination
of a field-effect transistor 533 and rectifier 531. Rectifiers 531 and 541
and their respective functions are described in greater detail below.
Second switch means 150 comprises field effect transistor 752, and
optionally a blocking rectifier 756 and a parasitic substrate diode 754.
These elements and their respective functions are described in greater
detail below. Each of transistors 533, 543, and 752 includes a control
terminal (gate) and two conduction terminals (source and drain). The gate
and source terminals of transistor 543 are coupled to control input port
103, the gate and source terminals of transistor 533 are coupled to
control input port 104, and the gate and source terminals of transistor
752 are coupled to control input port 105. Each of ports 103, 104, and 105
includes two terminals for receiving two signal lines for respective
coupling to these gate and source terminals. In a preferred embodiment of
the present invention, each of transistors 533, 543, and 752 comprises an
n-channel enhancement-type MOS field-effect transistor device. However, it
may be appreciated that each of switch means 130, 140 and 150 may
alternatively comprise other transistor devices, such as bipolar-junction
transistors (BJTs).
Rectifiers 531 and 541 enable their respective switch means 130 and 140 to
conduct under zero-voltage conditions before the corresponding switching
transistors 533 and 543 are rendered conductive by their respective
control signals. This enables the timing constraints for the signals
provided to switching transistors 533 and 543 to be relaxed in order to
achieve zero-voltage switching on switch means 130 and 140. More
specifically, transistor 543 may be rendered conductive a predetermined
time duration after transistor 533 is rendered non-conductive, and
transistor 533 may be rendered conductive a predetermined time duration
after transistor 543 is rendered non-conductive. As indicated above, the
time durations for segments 3/3', 4/4', and 6 shown in FIGS. 5-7 required
for zero-voltage switching are a function of the inductance L.sub.M of
magnetizing inductor 123, the capacitance C.sub.S of stray capacitor 132,
and the duty cycle. Thus, as the duty cycle varies in response to changes
in the input voltage or the load current demand, the time duration
(segment 6) between the opening of first switch means 130 and the closing
of primary switch means 140 needed for zero-voltage switching on primary
switch means 140 varies. Likewise, the time duration (segment 3/3' plus
segment 4/4') between the opening of primary switch means 140 and the
closing of first switch means 130 needed for zero-voltage switching on
first switch means 130 varies. Rectifiers 531 and 541 sense the zero
voltage switching condition across their corresponding switch means and
selectively conduct before the corresponding switching transistors are
rendered conductive, thereby automatically adjusting for variations in the
required durations of segments 3/3', 4/4', and 6 caused by variations in
the duty cycle and component values C.sub.S and L.sub.M.
More specifically, rectifier 541 is oriented such that it conducts a
forward current from the negative terminal of port 801 to the positive
terminal of port 802 (e.g., the cathode of rectifier 541 is coupled to
voltage node V.sub.1). Rectifier 531 is oriented such that it conducts a
forward current from voltage node V.sub.1 to storage capacitor 134 (e.g.,
the anode of rectifier 531 is coupled to voltage node V.sub.1). With
reference to timing diagrams 300 and 400 shown in FIGS. 6 and 7,
respectively, the following properties of operation for rectifiers 541 and
531 in converter 100 are apparent. Rectifier 541 conducts current from
supply input port 801 to the node V.sub.1 when the voltage between node
V.sub.1 and the negative terminal input port 801 becomes zero or negative.
In timing diagrams 300 and 400 shown in FIGS. 6 and 7, this condition
occurs when voltage V.sub.1 is driven to zero volts in segment 6, as shown
at points 311 and 313. Thus, the current in primary switch means 140 may
be conducted by rectifier 541 during the first part of segment 1 and,
possibly, segment 2/2', if switching transistor 543 is not rendered
conductive by its control signal. If transistor 543 remains open,
rectifier 541 will continue to conduct current during segments 1 and 6 as
long as the primary current, shown at graph 320 in FIG. 6 and 420 in FIG.
7, remains negative. Thus, the incorporation of rectifier 541 with switch
means 140 provides a relatively wide window in which transistor 543 may be
closed with respect to the opening of first switch means 130.
Likewise, rectifier 531 conducts current from node V.sub.1 to the positive
terminal of storage capacitor 134 when the voltage at node V.sub.1 is
equal to or greater than the voltage at the positive terminal of capacitor
134. In timing diagrams 300 and 400 shown in FIGS. 6 and 7, this condition
occurs when the voltage V.sub.1 is being driven to a high positive voltage
at the end of segments 4/4' by the magnetizing current I.sub.M, as shown
at 312 and 412, respectively. Thus, if switching transistor 533 is not
closed, the current in first switch means 130 may be conducted by
rectifier 531 during the first part of segment 5 since the direction of
the magnetizing current is the same as the conducting direction of
rectifier 531, as shown at graphs 330,340 and 430,440 in FIGS. 6 and 7. If
transistor 533 remains open during segment 5, rectifier 531 will continue
to conduct the current through first switch means 130 until the
magnetizing current I.sub.M reverses direction half-way through segment 5,
assuming steady-state operating conditions. The closing of transistor 533
may then be delayed until the midpoint of segment 5, assuming steady-state
operating conditions. Thus, the incorporation of rectifier 531 with switch
means 130 provides a relatively wide window in which switch means 533 may
be closed with respect to the opening of primary switch means 140.
In a broader viewpoint, rectifier 531 comprises means for detecting a
zero-voltage-switching condition across first switch means 130 and means
for initially conducting current through switch means 130 upon the
occurrence of a zero-voltage-switching condition across switch means 130.
Likewise, rectifier 541 comprises means for detecting a
zero-voltage-switching condition across primary switch means 140 and means
for initially conducting current through switch means 140 upon the
occurrence of a zero-voltage-switching condition across switch means 140.
As such, the delay in closing transistor 543 after the opening of
transistor 533 may be set to a predetermined, or fixed, value. The fixed
delay value is applicable to a predetermined range of duty-cycles and is
set to the longest time duration for segment 6 in the given range of
duty-cycles. Assuming a steady-state operation for converter 100, the
longest duration for segment 6 occurs for the lowest duty-cycle in the
given range. Operating at the lowest duty-cycle value in the range,
rectifier 541 begins to conduct at the same point that transistor 543 is
closed. As the duty-cycle increases with respect to the lowest value,
rectifier 541 conducts prior to the closing of transistor 543 and provides
for zero-voltage switching for primary switch means 540.
Likewise, the delay in closing transistor 533 after the opening of
transistor 543 may be set to a predetermined, or fixed, value for a
predetermined range of duty-cycles. In the case of sequence 230, shown in
FIG. 5, the fixed delay value is set to the longest time duration of the
sum of segments 3 and 4 in the given range of duty-cycles. Assuming a
steady-state operation for converter 100, the longest duration for the sum
of segments 3 and 4 occurs for the lowest duty-cycle in the given range.
In the case of sequence 240, shown in FIG. 5, the fixed delay value is set
to the longest time duration of segment 4' in the given range of
duty-cycles. Assuming a steady-state operation for converter 100, the
longest duration for segment 4' also occurs for the lowest duty-cycle in
the given range. Operating at the lowest duty-cycle value in the range,
rectifier 531 begins to conduct at the same point transistor 533 is
closed. As the duty-cycle increases with respect to the lowest value,
rectifier 531 conducts prior to the closing of transistor 533 and provides
for zero-voltage switching for first switch means 130. Such fixed delay
are employed by the embodiment of switch control means 820 shown in FIGS.
11A and 11B. The above discussion also applies to second converter 100',
which includes corresponding switching transistors 533' and 543' and
rectifiers 531' and 541'.
In second switch means 150, field effect transistor 752 may further include
a parasitic substrate diode, as indicated at 754 in FIG. 9. The inclusion
of parasitic substrate diode 754 depends upon the particular transistor
component used for transistor 752. If parasitic substrate diode 754 exists
as part of field effect transistor 752, an undesired current will be
conducted from the source terminal to the drain terminal when a negative
drain-to-source voltage is applied across transistor 752. To block this
undesired current, a blocking rectifier 756 coupled in series between the
drain terminal of transistor 752 and secondary winding 124 may be
included. As shown in FIG. 9, blocking rectifier 756 is oriented such that
current conducted by parasitic substrate diode 754 is prevented from
entering secondary winding 124.
If transistor 752 does not include parasitic substrate diode 754, blocking
rectifier 756 may be removed so as to directly couple the drain of
transistor 752 to secondary winding 124. In the case where blocking
rectifier 756 can and is eliminated and where transistor 752 comprises an
n-channel enhancement-type MOSFET, sufficient negative voltage should be
applied between the gate and source terminals of transistor 752 during the
non-conducting periods of second switch means 150 in order to maintain
transistor 752 in a non-conducting state. This is because, during the
non-conducting state (e.g., the converter's OFF-period), a negative
voltage of between -5 V to -15 V appears at the drain terminal of
transistor 752 when rectifier 756 is not present, which could cause
transistor 752 to inadvertently conduct if the gate voltage is not
sufficiently negative with respect to the negative voltage at the drain.
The control signal at port 105 may be generated to provide such a negative
voltage by circuits well known in the transistor drive art and a detailed
explanation thereof is not needed for one of ordinary skill in the art to
understand this aspect of the present invention and to make and use the
same. It may be appreciated that some transistor switching devices (e.g.,
BJT devices) may not require the above-described application of negative
voltage during the non-conductive state.
The operation of first converter 100, including the zero-voltage switching
conditions for switch means 140 and 130, is as described above with
reference to FIGS. 4-7. As described in greater detail below, control
means 820 provides the control signals to switch means 130, 140, and 150
for this operation. Likewise, in a preferred embodiment of the present
invention, the operation of second converter 100', including the
zero-voltage switching conditions for switch means 140' and 130', is as
described above with reference to FIGS. 4-7. As is also described in
greater detail below, control means 820 provides the control signals to
switch means 130', 140', and 150' for this operation. However, the
switching cycle for second converter 100' is substantially different from
the switching cycle for first converter 100. This difference is discussed
below in greater detail with reference to the operation of control means
820.
For each of converters 100 and 100', the symbolic notations V.sub.S,
V.sub.P, V.sub.1, V.sub.2, I.sub.P, I.sub.M, I.sub.1, I.sub.S2, I.sub.CR2,
I.sub.LOUT, I.sub.CC, and I.sub.CS shown in FIG. 9 have the same
designations and references as the corresponding notations shown in FIG.
4. The total current flow into first converter 100 is shown by the current
designation I.sub.IN at input terminal 101 and the total current flow into
first converter 100' is shown by the current designation I'.sub.IN at
input terminal 101'. The combined input currents of converters 100 and
100' is shown by the current designation I.sub.MAIN, which is equal to
I.sub.IN +I'.sub.IN. The output current of DAC converter 800 is equal to
the combined output current I.sub.LOUT and I'.sub.LOUT of converters 100
and 100', respectively, and is shown by the current designation I.sub.OUT
in FIG. 9.
Current I.sub.IN into converter 100 is equal to the sum of the magnetizing
current I.sub.M and the reflected current I.sub.1 of secondary winding 124
less the current I.sub.CC through clamp capacitor 134. In mathematical
terms, this is stated as I.sub.IN =I.sub.M +I.sub.1 -I.sub.CC. The
subtraction of, rather than the addition of, the clamp capacitor current
I.sub.CC in this embodiment of the present invention is due to the choice
of the reference direction of I.sub.CC into capacitor 134 from first
switch means 130. As discussed above, the magnetizing current I.sub.M is
first directed into clamp capacitor 134 during the first half of the OFF
period and then reversed and directed out of clamp capacitor 134 towards
node V.sub.1 during the second half of the OFF period. Because the current
path into clamp capacitor 134 is completed via a path which is in parallel
with primary winding 122, the magnetizing current I.sub.M and
clamp-capacitor current I.sub.CC cancel one another in the relationship
for I.sub.IN during the OFF period. Thus, the reversal of the magnetizing
current I.sub.M during the OFF period is not coupled to the input source
connected to input port 121. In a similar manner, the current I'.sub.IN
into second converter 100' is equal to the sum of the magnetizing current
I'.sub.M and the reflected current I'.sub.1 of secondary winding 124' less
the current I'.sub.CC through clamp capacitor 134': I'.sub.IN =I'.sub.M
+I'.sub.1 -I'.sub.CC.
Shown at 850 in FIG. 10 is a timing diagram of exemplary waveforms of key
voltages and currents of DAC converter 800, shown as a function of time.
Timing diagram 850 includes graphs 851-861 for showing the key voltages
and currents and is discussed more fully below. As an aid in practicing
the present invention, Table I lists the component values for a preferred
embodiment of DAC converter 800 which provides a 44 VDC-to-5 VDC converter
capable of providing a maximum output current of 40 A.
TABLE I
______________________________________
Input Capacitor 803
1.8uF, 100V
Output Capacitor 172
96uF, 10V (Two 47uF Capacitors)
Power Transformer 120
Turns ratio = 8:2 = 4:1 (pri:sec)
Transistor 140 IRFP250 (International Rectifier)
Transistor 130 IRFR220 (International Rectifier)
Transistor 150 IRFP054 (International Rectifier)
Capacitor 134 47nF, 200V
Sense Transformer 135
Turns ratio = 1:40 (pri:sec)
Rectifier 160 .about.0.5V forward Voltage
Inductor 170 1.8uH
Switching Frequency
500KHz
______________________________________
Having thus described the structure and operation of first converter 100
and second converter 100', the operation of control means 820 and the
coordination of operating converters 100 and 100' are now described. For
first converter 100, control means 820 includes an output control port 821
coupled to input control port 103 for providing a control signal to
transistor 543 of primary switch means 140, an output control port 822
coupled to input control port 104 for providing a control signal to
transistor 533 of first switch means 130, and an output control port 823
coupled to input control port 105 for providing a control signal to
transistor 752 of second switch means 150. Each of ports 821, 822, and 823
comprises two terminals which are coupled to respective terminals of input
ports 103, 104, and 105, respectively. In a preferred embodiment of DAC
converter 800, each of the control signals at ports 821, 822, and 823
appears as a voltage across the terminals of ports 821, 822, and 823,
respectively. For second converter 100', control means 820 further
includes an output control port 824 coupled to input control port 103' for
providing a control signal to transistor 543' of primary switch means
140', an output control port 825 coupled to input control port 104' for
providing a control signal to transistor 533' of first switch means 130',
and an output control port 826 coupled to input control port 105' for
providing a control signal to transistor 752' of second switch means 150'.
Each of ports 824, 825, and 826 comprises two terminals which are coupled
to respective terminals of input ports 103', 104', and 105', respectively.
In a preferred embodiment of DAC converter 800, each of the control
signals at ports 824, 825, and 826 appears as a voltage across the
terminals of ports 824, 825, and 826, respectively.
Control means 820 further includes a power port 830 coupled to input port
801 for receiving a source of power and an output monitoring port 827
coupled to output 802 for receiving the voltage provided to port 802.
Control means 820 further includes an input port 828 coupled to port 106
of first converter 100 for receiving a signal representative of the
primary current of first converter 100 and an input port 829 coupled to
port 106' of second converter 100' for receiving a signal representative
of the primary current of second converter 100'. As described below in
greater detail, control means 820 senses the output voltage at port 802
via port 827 and respective converter primary currents at ports 828 and
829, and, in response thereto, adjusts the duty-cycle of the control
signals present on control ports 821-826.
In operation, control means 820 coordinates the operation of first
converter 100 and second converter 100' such that the amount of ripple
current at input port 801 is minimized. This is done by providing
coordinated pulse-width modulated signals to ports 821-826. In one
embodiment of the present invention, the signals at ports 821-823 (bus
805) have a common switching cycle period T, or time duration T, which is
preferably substantially constant during operation of DAC converter 800.
Additionally, the signals at ports 824-826 (bus 806) have a common
switching cycle period T', or time duration T', which is substantially
equal to the duration of period T. The signals at ports 821-826, along
with key currents of converters 100 and 100', are shown as a function of
time by a timing diagram at 850 in FIG. 10. Specifically, the signals at
ports 821-826 are shown by graphs 851-856, respectively. In a preferred
embodiment of DAC converter 800, each of the signals at ports 821-826 has
a "resting" state of substantially zero volts and a "pulse" state of
substantially 10 volts, as seen in graphs 851-856, respectively. For each
of ports 821-826, the voltage appears across the port's terminals, and in
turn, across the gate and source terminals of the MOSFET transistor
coupled to the port. In the pulse state, a positive voltage is applied
between the gate and source terminals, with the gate terminal being more
positive.
The signals at ports 821 and 822 for primary switch means 140 and first
switch means 130, respectively, are complimentary to one another, and the
signals at ports 824 and 825 for primary switch means 140' and first
switch means 130', respectively, are complimentary to one another. For
example, the signal at port 821 is in its pulse state when the signal at
port 822 is in its resting state, and the signal at port 822 is in its
pulse state when the signal at port 821 is in its resting state. The
ensures that switch means 140 and 130 are switched opposite to one another
and that neither are conducting at the same time. The same applies to
switch means 140' and 130' of second converter 100'. In one embodiment of
DAC converter 800, the signals at ports 823 and 826 for second switch
means 150 and 150' are substantially delayed versions of the signals at
ports 821 and 824, respectively. This may be seen by comparing graphs 853
and 851, and graphs 856 and 854. This is according to the first
switching-event sequence 230 discussed above with reference to FIGS. 5 and
6. However, it may be appreciated that the second switching-event sequence
240 discussed above with reference to FIGS. 5 and 7 may also be used.
Each of the control signals at ports 821 and 824 (primary switch means)
starts its respective switching cycle with its pulse state, thereby being
in its pulse state during the beginning portion of its switching period T
and being in its resting state during the latter portion of its switching
period T (time duration T). Additionally, the control signal at port 824,
which is for primary switch means 140' of second converter 100', starts
its switching cycle substantially one-half switching-cycle period (T/2)
after the control signal at port 821 starts its switching cycle, the
signal at port 821 being for primary switch means 140 of first converter
100. Thus, converters 100 and 100' operate substantially 180 degrees out
of phase. This offset is shown at the beginning of graph 854 at 865.
Also in a preferred embodiment of the present invention, the duty cycle of
the control signals at ports 821 and 824 is nominally set to a value of
substantially 0.50 (50%). Additionally, the control signals at ports 821
and 824 are coordinated to have substantially the same duty cycle during
operation in their respective switching periods T and T'. The pulse
duration for each of the control signals at ports 821 and 824 (primary
switch means 140 and 140', respectively) is equal to the duty cycle
multiplied by the corresponding switching period, e.g., (duty cycle)*T.
The pulse duration for each of the control signals at ports 822 and 825
(first switch means 130 and 130', respectively) is equal to the fraction
1.0 less the duty cycle, the difference multiplied by the corresponding
switching period, e.g., (1-(duty cycle))*T.
In a further embodiment of the present invention, the duty cycle for the
control signals at ports 821 and 824 (for the primary switch means) is
targeted for a nominal operating value of substantially 0.50 (50%) and an
operating range of 0.30 (30%) to 0.70 (70%). This allows a substantial
amount of ripple-current reduction during nominal operation at 0.50 (50%)
and allows for compensating over- and under-voltages at the input by
moving the duty-cycle towards 0.30 (30%) and 0.70 (70%), respectively,
while still maintaining good input and output ripple current
characteristics. Examples of the input and output ripple reduction as a
function of duty-cycle are provided below.
In a preferred embodiment of the present invention, the duty cycle for the
control signals at ports 821 and 824 (for the primary switch means) ranges
between 0.40 and 0.65 about a nominal value of 0.50. As such, the
corresponding, complimentary duty cycle for the control signals at ports
822 and 825 (for the first switch means) ranges between 0.60 and 0.35. As
is known in the forward converter art, the duty cycle for the primary
switch means (ports 821 and 824) is adjusted to regulate the voltage at
output port 802 and to compensate for variations in the voltage at input
port 801. If the output voltage decreases from its targeted nominal value,
control means 820 increases the duty cycle to provide more current to
output capacitor 172 and the load at port 802 in order to increase the
output voltage. If the output voltage increases above its nominal value,
the duty cycle is decreased to provide less current. If the input voltage
decreases from its nominal value, less voltage is provided to the
secondary circuits of converters 100 and 100' during their respective ON
periods and, hence, less energy is coupled to the output by the secondary
circuits. To compensate for the energy loss, control means 820 increases
the duration of the ON periods to increase the amount of energy coupled so
as to maintain the output voltage at its nominal value.
The steady-state output voltage for each of converters 100 and 100' under
nominal current loads is related to the input voltage and duty-cycle by
the following equation:
##EQU2##
where V.sub.OUT is the output voltage, V.sub.F is the voltage losses in
the secondary circuit (mainly from the secondary-side rectifiers 160,
160', 756, and 756', and transistors 752 and 752'), V.sub.IN is the input
voltage, and N.sub.TURNS is the turns ratio of the power transformer, such
as 120 and 120' (number of primary turns divided by the number of
secondary turns). In the present invention, the duty cycle is varied in
response to changes in the input voltage, so as to maintain the output
voltage V.sub.OUT at a targeted nominal value.
To realized a 50% duty cycle operation for converters 100 and 100' at the
nominal output voltage V.sub.OUT,NOM, the ratio of the nominal input
voltage V.sub.IN,NOM to the transformer turns ratio N.sub.TURNS is set as
follows:
##EQU3##
Generally, the nominal input voltage V.sub.IN,NOM is a given parameter,
i.e., a constraint, leaving the transformer turns ratio N.sub.TURNS as the
parameter to be chosen to satisfy equation (4). For example, if DAC
converter 800 is to have a nominal input voltage of V.sub.IN,NOM =48 V
with a nominal output voltage of V.sub.OUT,NOM =5 V and a typical voltage
loss V.sub.F of 0.5 V, the transformer turns ratio N.sub.TURNS for each
converter 100 and 100' would be substantially set to 4.36:1, or 13:3, to
satisfy equation (4). In this way, the transformer turns ratio may be used
to effectively set the duty cycle for each of converters 100 and 100' to
substantially 50% (0.50) for operation at the given nominal input voltage
and nominal output voltage. For a nominal input voltage of 44 V and a
nominal output voltage of 5 V, N.sub.TURNS would be substantially 4:1, or
an integer multiple thereof. As described below in greater detail, control
means 820 operates to keep the output voltage close to the nominal output
voltage. Thus, when the input voltage is at its nominal value, control
means 820 operates to keep the duty cycle for each of converters 100 and
100' near 50%.
For generating the control signals at port 821-826, control means 820
further comprises a pulse width modulation (PWM) means 910, as shown in
FIG. 8., for generating a first duty-cycle signal S.sub.1 for controlling
first converter 100 and a second duty-cycle signal S.sub.2 for controlling
second converter 100'. Each of the first and second duty-cycle signals has
a first state and a second state, and each alternates in time between its
respective first and second states. Additionally, the second duty-cycle
signal is phase-shifted in time by substantially 180 degrees from the
first duty-cycle signal. In a preferred embodiment of the present
invention and as described below in greater detail, the first duty-cycle
signal is coupled to port 821 (for primary switch means 140 of converter
100) substantially unchanged and the second duty-cycle signal is coupled
to port 824 (for primary switch means 140' of converter 100')
substantially unchanged. Each of the duty-cycle signals is generated as a
function of the measured output voltage and the targeted nominal output
voltage such that the output voltage is regulated to be within a specified
range around the nominal output voltage. An exemplary range is .+-.10 mV
around a nominal output voltage of 5 V. In a preferred embodiment of DAC
converter 800, each of the duty-cycle signals is also a function of the
primary input current of its respective converter (current-mode control).
In turn, the primary input current waveform is a function of the input
voltage, as described in greater detail below, thereby making each of the
duty cycle signals a function of the input voltage. As such, with the
transformer turns ratio N.sub.TURNS for each converter 100 and 100' set
for 50% duty-cycle operation as described above, each of the duty cycle
signals is substantially 0.50 (50%) when the input and output voltages are
at their respective nominal values.
As indicated above, first switch means 130 and 130' are operated in a
complimentary manner with respect to their corresponding primary switch
means 140 and 140', respectively. For this, control means 820 further
comprises a first control means 970, as shown in FIG. 8, which is
responsive to the first duty-cycle signal and is for operating primary
switch means 140 and first switch means 130 of first converter 100 in a
complimentary manner. Specifically, first control means 970 generates a
pulse state at port 821 to cause primary switch means 140 to close
substantially when the first duty-cycle signal enters its respective first
state, and generates a resting state at port 821 to cause primary switch
means 140 to open substantially when the first duty-cycle signal enters
its respective second state. Additionally, first control means 970
generates a resting state at port 822 to cause first switch means 130 to
open substantially when the first duty-cycle signal enters its respective
first state, and generates a pulsed state at port 822 to close
substantially when the first duty-cycle signal enters its respective
second state. In a preferred embodiment of the present invention, the
pulse state for the signals at ports 821 and 822 are non-overlapping such
that switch means 130 and 140 are not rendered conductive at the same
time. In an embodiment described below, the pulsed states of the signals
at ports 821 and 822 are separated by predetermined time durations. In one
embodiment of the present invention, first control means 970 is also for
operating second switch means 150 and generates a signal at port 823 to
cause second switch means 150 to close a time duration after primary
switch means 140 closes and to open substantially when primary switch
means 140 opens. In this embodiment, as described in greater detail below
with reference to FIGS. 11 and 12, the first control means causes second
switch means 150 to open a short time duration after primary switch means
140 opens.
In a similar manner, control means 820 further comprises a second control
means 999, as shown in FIG. 8, which is responsive to the second
duty-cycle signal and is for operating primary switch means 140' and first
switch means 130' of second converter 100' in a complimentary manner.
Specifically, second control means 999 generates a pulse state at port 824
to cause primary switch means 140' to close substantially when the second
duty-cycle signal enters its respective first state, and generates a
resting state at port 824 to cause primary switch means 140' to open
substantially when the second duty-cycle signal enters its respective
second state. Additionally, second control means 999 generates a resting
state for the signal at port 825 to cause first switch means 130' to open
substantially when the second duty-cycle signal enters its respective
first state, and generates a pulse state at port 825 to close
substantially when the second duty-cycle signal enters its respective
second state. In a preferred embodiment of the present invention, the
signals at ports 824 and 825 are non-overlapping such that switch means
140' and 130' are not rendered conductive at the same time. In an
embodiment described below, the pulsed states of the signals at ports 824
and 825 are separated by predetermined time durations. In one embodiment
of the present invention, the second control means is also for operating
second switch means 150' and generates a signal at port 826 to cause
second switch means 150' to close a time duration after primary switch
means 140' closes and to open substantially when primary switch means 140'
opens. In this embodiment, as described in greater detail below with
reference FIGS. 11 and 12, the second control means causes second switch
means 150' to open a short time duration after primary switch means 140'
opens.
In a further preferred embodiment of the present invention, PWM means 910
generates the duty cycles signals S.sub.1 and S.sub.2 such that the first
state of each duty-cycle signal is the pulse state and the second state is
the resting state. Additionally, each of the duty-cycle signals S.sub.1
and S.sub.2 further comprises a plurality of consecutive switching
periods, each said switching period have a substantially constant time
duration T. Each switching period begins when its corresponding duty-cycle
signal enters its respective first state from its respective state,
continues when its corresponding duty-cycle signal enters its respective
second state from its respective first state, and ends when its
corresponding duty-cycle signal next enters its respective first state
from its respective second state. Further, the consecutive switching
periods of the second duty-cycle signal start substantially one-half of
the time duration T after the consecutive switching periods of the first
duty-cycle signal start. Stated in another way, the second duty-cycle
signal enters its respective first (pulse) state substantially T/2 after
the first duty-cycle signal enters its respective first (pulse) state. In
this way, PWM means 910 generates the duty cycle signals such that the
second duty-cycle signal is phase-shifted by substantially 180 degrees
from the first duty-cycle signal.
In a preferred embodiment of the present invention, the duration of the
first (pulse) states of the duty cycle signals are substantially the same.
In other words, the first and second duty-cycle signals are coordinated to
have substantially the same duty-cycle value by PWM means 910. This aids
in having first converter 100 and second converter 100' equally provide
power to output port 802.
The reduction of the input ripple current at input port 801 by the
coordination of control signals at ports 821-826 is explained with
reference to timing graphs 857-860 shown in FIG. 10. The following
parameters are taken; a duty cycle of 0.50 (50%); a 4:1 turns ratio for
power transformers 120 and 120'; a nominal input voltage of 44 V, a
nominal output voltage of 5 V, 40 A output load current (maximum), 20 A
for each of converters 100 and 100'; a nominal output ripple current of 3
A in each of converters 100 and 100'; and a nominal magnetizing ripple
current of 0.25 A for each of converters 100 and 100'. Graph 857 shows the
magnetizing currents I.sub.M and I'.sub.M of converters 100 and 100',
respectively; graph 858 shows the clamp currents I.sub.CC and I'.sub.CC of
converters 100 and 100', respectively; graph 859 shows the output currents
I.sub.LOUT and I'.sub.LOUT of converters 100 and 100', respectively; and
graph 860 shows the primary input currents I.sub.IN and I'.sub.IN of
converters 100 and 100', respectively. The currents I.sub.M, I.sub.CC,
I.sub.LOUT, and I.sub.IN for first converter 100 are shown by solid lines
in graphs 857-860, and the currents I'.sub.M, I'.sub.CC, I'.sub.LOUT, and
I'.sub.IN for second converter 100' are shown by dashed lines in graphs
857-860.
As seen in graph 857, the magnetizing currents I.sub.M and I'.sub.M rise
during the ON periods of their respective converters and fall during the
OFF periods of their respective converters. In a preferred embodiment of
the present invention, both magnetizing currents are centered around zero
amperes. As indicated above, each of the magnetizing currents I.sub.M and
I'.sub.M is directed into its corresponding clamp capacitor 134 and 134',
respectively, during the OFF period of its converter. This may be seen in
graph 858 where I.sub.CC is substantially equal to I.sub.M during the OFF
period of first converter 100 and where I'.sub.CC is substantially equal
to I'.sub.M during the OFF period of second converter 100'. As indicated
above, currents I.sub.CC and I'.sub.CC are substantially zero during the
ON periods of their respective converters because first switch means 130
and 130' are not conducting during the respective OFF periods.
Each of the inductor currents I.sub.LOUT and I'.sub.LOUT, as shown in graph
859, rises at a substantially uniform rate during its converter's ON
period and falls at a substantially uniform rate during its converter's
OFF period. The rates of increase and decrease (i.e., slopes) of these
currents have substantially the same magnitude, which is due to the
duty-cycle being at substantially 0.50 (50%). Each of current I.sub.LOUT
and I'.sub.LOUT are centered about an average D.C. value, which represents
the average curent delivered to the output by converters 100 and 100',
respectively. Due to the substantially 180 degree phase difference in the
operation of first converter 100 and second converter 100', current
I'.sub.LOUT decreases when current I.sub.LOUT increases, and current
I'.sub.LOUT increases when current I.sub.LOUT decreases. Thus, the ripple
in currents I.sub.LOUT and I'.sub.LOUT are complimentary to one another
and, when combined at output port 802, cancel one another to leave
substantially no ripple current at output port 802 for a duty cycle of
0.50 (50%).
The output current at port 802 is designated as current I.sub.OUT and is
shown in FIG. 9. The waveform characteristic of I.sub.OUT is shown in
graph 859 of FIG. 10 as I.sub.OUT /2 (dotted line) to better shown the
cancellation of the ripple components in currents I.sub.LOUT and
I'.sub.LOUT. With this reduction in the output ripple current, the
capacitance value and displacement volume of output capacitor 172 may be
significantly reduced over that required by a single active clamp forward
converter providing the same power capability. (As a comparison, such a
single converter would have an output ripple current twice that of current
I.sub.LOUT, or that of current I'.sub.LOUT.) In the best case of a 50%
duty cycle, no output capacitance would be needed to absorb the ripple
current. However, as indicated above, the duty-cycle is often varied to
compensate for changes in input voltage and output current demand. As the
duty cycle of DAC converter 800 is changed from a value of 0.50 (50%), the
amount of ripple current increases in proportion to the magnitude of the
deviation away from 0.50, i.e., .alpha. .parallel.0.50-duty
cycle.parallel.. A more detailed analysis of the output ripple current is
given below after the input current I.sub.IN and I'.sub.IN are discussed.
The primary input currents I.sub.IN and I'.sub.IN to converters 100 and
100', respectively, are shown in graph 860 of FIG. 10. Each of the input
currents I.sub.IN and I'.sub.IN rises at a substantially linear rate
during the ON period of its corresponding converter, and drops to zero
during the OFF period of its corresponding converter. As indicated above,
the current I.sub.IN during the ON period of converter 100 is equal to the
sum of the magnetizing current I.sub.M and to the reflected secondary
current I.sub.1, which is substantially equal to I.sub.LOUT /4 since the
turns ratio of transformer 120 is taken as 4:1. Thus, these components in
graphs 857 and 859 add to give I.sub.IN in graph 860. During the OFF
period, as indicated above, the magnetizing current I.sub.M is directed
into clamp capacitor 134, and therefore is not reflected in I.sub.IN.
Additionally, the secondary current flow during the OFF period is
substantially supported by rectifier 160 (CR2), and therefore is not
reflected in I.sub.IN. Thus, I.sub.IN is substantially zero during the OFF
periods, as shown in graph 860. A similar analysis applies to primary
input current I'.sub.IN for second converter 100' in relation to its
waveform shown in graph 860.
The summation of the input currents I.sub.IN and I'.sub.IN occurs at input
port 801, and is designated as I.sub.MAIN in FIG. 9 and graph 860 of FIG.
10 (dotted line). As can be seen in graph 860, the current I.sub.MAIN
follows the alternating "ripples" of I.sub.IN and I'.sub.IN. I.sub.MAIN
comprises a DC component and an AC ripple component, which includes the
ripple currents from the magnetizing currents and the ripple currents of
the reflected secondary currents. In the exemplary embodiment indicated
above, each magnetizing ripple current has a peak-to-peak variation of
.about.0.25 A and each of the secondary ripple currents has a peak-to-peak
variation of .about.3 A. When reflected to the primary circuit with a 4:1
turns ratio, the peak-to-peak variation of each reflected secondary ripple
current is .about.0.75 A. Thus, the peak-to-peak variation of the ripple
current of I.sub.MAIN in this exemplary embodiment is approximately 1.0 A
(0.25 A+0.75 A). This ripple current is approximately 25% of the maximum
DC component of I.sub.MAIN, which is .about.5 A. The AC ripple current of
I.sub.MAIN is shown in greater detail by a timing diagram 1300 in FIG. 15,
which shows at a graph 1310 the input ripple currents of a single active
clamp forward converter, of a DAC converter 800 comprising converters 100,
and another of a DAC converter 800 according to the present invention
discussed in greater detail below. The AC ripple current of I.sub.MAIN is
shown at 1314 in FIG. 15.
As indicated above, the active clamp comprising capacitor 134 and first
switch means 130 may also be coupled in parallel with the secondary
winding of transformer 120. Such a converter suitable for use in DAC
converter 800 is shown at 1100 in FIG. 13 as a second embodiment of the
active-clamp forward converter according to the present invention.
Converter 1100 comprises the same elements as those of converter 100 shown
in FIG. 9. The reference number of each element of converter 1100
comprises the reference number of the corresponding element of converter
100 plus an appended designation "a". The elements of converter 1100 are
coupled to one another is the same manner as the elements of converter 100
shown in FIG. 9 with the exception that the series combination of
capacitor 134a and first switch means 130a is coupled in parallel with the
secondary winding 124a, before second switch means 150a. (It may also be
appreciated that the active clamp may also be coupled in parallel to a
third winding of power transformer 120, which would be dedicated for the
active clamp.) A DAC converter 800 comprising converters 100a has
input-ripple-current and output-ripple-current characteristics
substantially similar to those of a DAC converter 800 which comprises
comverters 100.
The active clamp comprising capacitor 134 and first switch means 130 may
also be coupled in parallel with primary switch means 140. Such a
converter suitable for use in DAC converter 800 is shown at 1200 in FIG.
14 as a third embodiment of the active-clamp forward converter according
to the present invention. Converter 1200 comprises the same elements as
those of converter 100 shown in FIG. 9. The reference number of each
element of converter 1200 comprises the reference number of the
corresponding element of converter 100 plus an appended designation "b".
The elements of converter 1200 are coupled to one another is the same
manner as the elements of converter 100 shown in FIG. 9 with the exception
that the series combination of capacitor 134b and first switch means 130b
is coupled in parallel with primary switch means 140b. A DAC converter 800
comprising converters 100b has an output-ripple-current characteristic
substantially similar to that of a DAC converter 800 which comprises
converters 100.
However, a DAC converter 800 which comprises a first converter 100b and a
second converter 100b' has an input-ripple-current characteristic in
comparison to that of a DAC converter 800 which comprises two converters
100. This is because the magnetizing currents of the converters 100b and
100b' are "counter-balanced" against one another. The parallel coupling of
the active clamp with the primary switch means (140b, 140b') allows the
magnetizing current (I.sub.M, I'.sub.M) during the OFF period of its
respective converter (100b, 100b') to be coupled through to input port 801
and capacitor 803 and, therefore, to show up in the converter input
current (I*.sub.IN, I*.sub.IN ') during the OFF period. The input currents
of two such converters 100b and 100b' are designated as I*.sub.IN and
I*.sub.IN ', respectively, and are shown in a graph 861 in FIG. 10. As
indicated above with regard to forward converter 100, the primary input
current I.sub.IN of converter 100 is equal to the sum of the magnetizing
current I.sub.M and the reflected secondary current I.sub.1 less the clamp
capacitor current I.sub.CC , of which the currents I.sub.M and I.sub.CC
substantially canceled one another in the formation of I.sub.IN during the
OFF period of converter 100. Therefore, in converter 100 shown in FIG. 9,
the current I.sub.IN is substantially zero during the OFF period (as
indicated above, the reflected secondary current is zero during this
time). In converter 100b, however, the primary input current I*.sub.IN is
equal to the sum of the magnetizing current I.sub.M and the reflected
secondary current I.sub.1. The current I.sub.CC does not form part of the
mathematical summation of the input current I*.sub.IN and, as such, the
currents I.sub.M and I.sub.CC of converter 100b (and 100b') do not cancel
one another in the formation of I*.sub.IN during the OFF period of
converter 100b (and 100b'). Therefore, the current I*.sub.IN is
substantially equal to the current I.sub.M during the OFF period of
converter 100b (and 100b').
When combined at input port 801 of a DAC converter 800, the magnetizing
current component of I*.sub.IN during the OFF period of the first
converter 100b substantially cancels the magnetizing current component of
I*.sub.IN ' during the ON period of the second converter 100b and,
vice-versa, the magnetizing current component of I*.sub.IN ' during the
OFF period of the second converter 100b' substantially cancels the
magnetizing current component of I*.sub.IN ' during the ON period of the
first converter 100b. The cancellation may be seen in the waveform of the
combined currents I*.sub.MAIN (=I*.sub.IN +I*.sub.IN '), which is shown in
graph 861 of FIG. 10 (dotted line). As can be seen by comparing graphs 861
and 860, the ripple of input current I*.sub.MAIN is substantially less
than that of the current I.sub.MAIN of DAC converter 800 comprising
converters 100. The ripple current that remains in current I*.sub.MAIN is
due to the ripple current present in the inductor currents I.sub.LOUT and
I.sub.LOUT ', as reflected to the primary circuit. As discussed above, the
inductor currents I.sub.LOUT and I.sub.LOUT ' are reflected to the primary
circuit substantially only during the ON periods of their respective
converters. Thus, the rising edge ripples in currents I.sub.LOUT and
I.sub.LOUT ' are propagated to I*.sub.MAIN. Using the component values
listed in TABLE I and a peak-to-peak value of 3 A for the output ripple
current of each converter 100b and 100b', the peak-to-peak value of the
output ripple current as reflected in I*.sub.MAIN of the DAC converter is
.about.0.75 A. This is approximately 0.25 A less than the ripple current
of I.sub.MAIN for a DAC converter comprising converters 100, which is 1.0
A, and represents a 25% reduction in the amount of input ripple current.
The AC ripple current of I*.sub.MAIN is shown in greater detail at 1316 in
graph 1310 of timing diagram 1300, which is shown in FIG. 15.
The ripple current for a single active clamp forward converter having the
same output power capability as DAC converter 800 is shown at 1312 in
graph 1310 of timing diagram 1300. This characteristic may be obtained by
multiplying the waveform of I.sub.IN shown in graph 860 (FIG. 10) by 2 (to
equalize output power capabilities) and centering the resulting waveform
above an average value. The peak-to-peak variation for the input ripple
current of the single active clamp forward converter shown at 1312 is
approximately 11.25 A, which is approximately eleven times the
peak-to-peak variation for I.sub.MAIN of DAC converter 800, which is
.about.1.0 A.
Timing diagram 1300 also shows at a graph 1320 the ripple voltages that
result from the input ripple currents being applied to a fixed reference
capacitor of 1 Farad. To obtain the waveforms for the corresponding ripple
voltages, the ripple currents are integrated as a function time over a
switching period according to the formula: V=(1/C)*.intg.I dt, C is taken
as 1 Farad for a ready basis of comparison. The ripple voltage for the
single active clamp converter is shown at 1322 in FIG. 15, and the ripple
voltage for DAC converter 800 comprising converters 100 and 100' is shown
at 1324. (The ripple voltage shown at 1322 has been scaled by a factor of
one-half so that it may be shown with the other voltages and currents of
FIG. 15.) The ripple voltage shown at 1324 has a peak-to-peak value which
is roughly forty times smaller than the ripple voltage shown at 1322. This
is because the peak-to-peak value for the ripple current of DAC converter
800 is roughly 11 times smaller than that for the single active clamp
converter, because the ripple current for DAC converter 800 is triangular
whereas the ripple current of the single converter is substantially
rectangular, and because the repetition period for the ripple current of
DAC converter 800 (i.e., T/2) is one-half that of the single active clamp
converter (i.e., T). This means that DAC converter 800 comprising
converters 100 and 100' may have, for the same amount of power output
capability, an input capacitance which is 2.5% of that required for a
single active clamp converter or, conversely, has 2.5% of the input ripple
voltage for the same amount of input capacitance.
The input ripple current component of I*.sub.MAIN for a DAC converter 800
comprising converters 100b and 100b' is shown at 1316 in FIG. 15. As seen,
I*.sub.MAIN has a peak-to-peak variation which is roughly 75% that of
I.sub.MAIN shown at 1314. The corresponding input ripple voltage is shown
at 1326 in FIG. 15, and is calculated in the same manner as that for
I.sub.MAIN. The peak-to-peak variation in the input ripple voltage for a
DAC converter 800 comprising converters 100b and 100b' is roughly 75% that
of a DAC converter 800 comprising converters 100 and 100', and roughly
1/50.sup.th that of the single active clamp forward converter. This means
that a DAC converter 800 converter comprising converters 100b and 100b'
may have, for the same amount of power output capability, an input
capacitance which is 2% of that required for a single active clamp
converter or, conversely, may have 2% of the output ripple voltage for the
same amount of input capacitance.
As indicated above, the output ripple current for DAC converter 800 is
essentially zero at a duty cycle of 0.50 (50%). This would be an
"infinite" improvement over that of the single active clamp forward
converter. However, as indicated above, a finite output ripple current is
produced when the duty cycle varies away from 0.50. An analysis of the
output ripple current shows that, in the range of duty cycles from 0.3 to
0.7, the peak-to-peak variation in the ripple current (I.sub.RPP) divided
by the corresponding variation in a single active clamp forward converter
providing the same power capability is given approximately as
(0.88*D+0.25*D.sup.0.5), where D=.parallel.0.50-duty.sub.--
cycle.parallel.. In a preferred embodiment of DAC converter 800, the duty
cycle ranges between 0.40 (D=0.10) and 0.65 (D=0.15). Thus, taking the
larger value of D at 0.15 (duty cycle=0.65), the maximum output ripple
current of DAC converter 800 is only 23% of the output ripple current of a
single active clamp forward converter providing the same output power
capability.
A further analysis shows that the repetition period of the output ripple
current of DAC converter 800 is half that of the single active clamp
converter. The combination of halving the repetition period and of
reducing the peak-to-peak variation by approximately a factor of 4.3
reduces the amount of ripple voltage at the output by approximately a
factor of 8.6. As indicated above with regard to the input ripple current,
the amount of output ripple voltage induced by the output ripple current
can be calculated by integrating the capacitance formula I.sub.RPP
=C.sub.OUT dV.sub.RPP /dt to solve for V.sub.RPP. Rather than performing
the integration, however, the magnitude of the differential V.sub.RPP can
be estimated with the formula: dV.sub.RPP =1/C.sub.OUT *I.sub.RPP *dt.
Thus, reducing the rippled current I.sub.RPP by a factor of 4.3 reduces
dV.sub.RPP by approximately a factor of 4.3, and reducing the integration
time dt by a factor of 2 (by halving the repetition period) further
reduces dV.sub.RPP by approximately a factor of 2, for a total reduction
of approximately 8.6. Thus, for a given output capacitance value, the
"noise" at the output of DAC converter 800 is roughly an order of
magnitude less than that of the single active clamp forward converter.
Exemplary embodiments of PWM means 910, first control means 970, and second
control means 999 of control means 820 shown in FIG. 9 are now described
in greater detail with reference to a more detailed schematic diagram of
control means 820 shown in FIGS. 11A and 11B. Shown in FIGS. 11A and 11B
are output control ports 821-826 for the switch means of converters 100
and 100', power port 830, and output monitoring port 827 for receiving a
signal related to the voltage provided to output port 802 of DAC converter
800. Also shown in FIGS. 11A and 11B are input ports 828 and 829, each for
receiving signals representative of the primary current of converters 100
and 100', respectively.
PWM means 910 is for generating a first duty-cycle signal S.sub.1 at a port
911 for controlling first converter 100, shown in FIG. 9, and for
generating a second duty-cycle signal S.sub.2 at a port 912 for
controlling second converter 100'. First control means 970 is responsive
to the first duty-cycle signal at port 911 and is for operating primary
switch means 140 and first switch means 130 of first converter 100 in a
complementary manner, as describe above. First control means 970 is also
for operating second switch means 150, causing it to close a time duration
after primary switch means 140 closes and to open a short time duration
after primary switch means 140 opens. First control means 970 provides a
control signal for primary switch means 140 at control port 821, a control
signal for first switch means 130 at control port 822, and a control
signal for second switch means 150 at control port 823. Likewise, second
control means 999 is responsive to the second duty-cycle signal at port
912 and is for operating primary switch means 140' and first switch means
130' of second converter 100' in a complementary manner, as described
above. Second control means 999 is also for operating second switch means
150', causing it to close a time duration after primary switch means 140'
closes and to open a short time duration after primary switch means 140'
opens. Second control means 999 provides a control signal for primary
switch means 140' at control port 824, a control signal for first switch
means 130' at control port 825, and a control signal for second switch
means 150' at control port 826.
PWM means 910 comprises a dual clock generator 920, a first pulse width
controller 950 and associate circuitry, and a second pulse width
controller 960 and associated circuitry. In a preferred embodiment of
control means 820, each of controllers 950 and 960 comprises an AS3843
current mode PWM controller manufactured by Astec Semiconductor. Dual
clock generator 920 provides two anti-phase clock signals at two terminals
921 and 922, respectively, which are for driving the two controllers 950
and 960. The anti-phase clock signals are phase shifted with respect to
one anther by substantially 180.degree.. Controller 950 is responsive to
the first anti-phase clock at terminal 921, to the signal at port 827,
which is related to the output voltage of converter 800, and to the signal
at port 828, which is representative of the primary current of converter
100. In response to these signals, controller 950 generates the first
duty-cycle signal at port 911 so as to maintain the output voltage of DAC
converter 800 near its nominal value. Specifically, controller 950 begins
the pulse of the first duty cycle signal in response to the clock signal
at terminal 921 and ends the pulse in response to the signals
representative of the output voltage and primary current of converter 100.
Controller 960 is responsive to the second anti-phase clock signal at
terminal 922, to the signal at port 829, which is representative of the
primary current of converter 100', and to a signal generated by controller
950 (provided by the "COMP" pin), which is described in greater detail
below. In response to these signals, controller 960 generates the second
duty-cycle signal at port 912. Specifically, controller 960 begins the
pulse of the second duty cycle signal in response to the clock signal at
terminal 922 and ends the pulse in response to the signal provided by
controller 950 and to the signal representative of the primary current of
converter 100'. Since the pulse state of each of the duty cycle signals is
initiated in response to its respective clock signal and since the clock
signals are phase shifted with respect to one another by substantially
180.degree. (i.e., in anti-phase), the first and second duty cycle signals
are phase shifted with respect to one another by substantially
180.degree..
PWM means 910 further includes an internal power supply 915 for providing a
low level supply of approximately 12 volts at a terminal 918 to power the
various components of PWM means 910 and control means 970 and 999.
Internal supply 915 comprises an input port 916 coupled to port 830 for
receiving a source of power from the input supply, a ground terminal 917
for providing a ground reference for control means 820, and a power
terminal 918 for supplying 12 volts to the various components of means
910, 970, and 999. Internal supply 915 preferably comprises its own power
transformer having a primary winding coupled to the input supply at port
830 and a secondary winding coupled between terminals 917 and 918. This
power transformer facilitates a ready isolation of the secondary ground at
terminal 917 from the input supply at port 830. Ground terminal 917 is
preferably coupled to the secondary-side grounds of converters 100 and
100' and to the secondary output ground at output port 802 of DAC
converter 800, as shown in FIG. 9. As such, the secondary outputs of
converters 100 and 100' and the circuitry of control means 820 share the
same ground. This allows control means 820 to directly sense the voltage
at output port 802 of DAC converter 800 without the need for an
optocoupler device. Internal supply 915 need only supply the amount of
current required by the components of control means 820, which is
typically less than 0.1 A.
The two anti-phase clocks generated at terminals 921 and 922 by dual clock
generator 920 are designated as .PHI..sub.1 and .PHI..sub.2, respectively.
Each of the clock signals .PHI..sub.1 and .PHI..sub.2 preferably has a
first logic state of approximately zero volts and a second logic state of
preferably approximately 2.5 volts, and each of the clock signals
alternates between its respective first and second states. Additionally,
each of the clock signals .PHI..sub.1 and .PHI..sub.2 has a switching
cycle period T of approximately 2.0 microseconds and a duty-cycle of 0.75
(75%), with the first state being the pulsed state. Also, as indicated
above, clock signals .PHI..sub.1 and .PHI..sub.2 are phase shifted with
respect to one another by substantially 180.degree., or equivalently,
offset in time by one-half of a switching cycle period (T/2=1.0
microseconds). Each of the clock signals .PHI..sub.1 and .PHI..sub.2 are
shown as a function of time by a timing diagram 1000 in FIG. 12 in
respective timing graphs 1005 and 1006 of timing diagram 1000. Timing
diagram 1000 also shows a number of additional signals generated by PWM
means 910 which will aid in explaining the generation of .PHI..sub.1,
.PHI..sub.2, and the first and second duty cycle signals at ports 911 and
912. These additional signals are shown at timing graphs 1001-1004, and
1007-1012.
For generating clock signals .PHI..sub.1 and .PHI..sub.2, dual clock
generator 920 comprises a conventional crystal oscillator for generating a
base clock signal, and a D-type flip-flop 930 which is responsive to the
base clock and is configured as a toggle flip-flop. Flip-flop 930 includes
a clock port "CK" which is coupled to the base clock signal, a data input
port "D", an output "Q" and a complementary output "Q/" coupled back to
the D input. Flip-flop 930 provides two anti-phase signals at the Q and Q/
outputs, which switch at half the switching frequency of the base clock
signal. Dual clock generator further comprises two logic OR gates for
digitally OR-ing the base clock signal with each of the anti-phase signals
of flip-flop 930. One OR gate includes two rectifiers 931-932 and two
resistors 935-936 and the other OR gate includes two rectifiers 933-934
and two resistors 937-938.
The crystal oscillator includes a crystal 926, a first inverter 924, and a
second inverter 929. Crystal 926 is preferably a 1.0 MHz crystal and is
coupled between the input and output of first inverter 924, which is
preferably a Schmitt-trigger CMOS inverter. A 1 M-ohm resistor 925 is also
coupled between the input and output of inverter 924 and is for biasing
inverter 924 near the middle of its gain curve characteristic.
Additionally, a capacitor 927 of approximately 33 pF is coupled to the
input of inverter 924 and a second capacitor 928 of approximately 33 pF is
coupled to the output of oscillator 924. With this configuration, a
substantially square wave is produced at the output of inverter 924. This
output is designated as "OSC" in FIG. 11A and is shown in a timing graph
1001 in FIG. 12. Second inverter 929 serves to isolate the crystal
oscillator from the remaining circuitry of clock generator 920. The input
of second inverter 929 is coupled to the "OSC" output. The output of
second inverter 929 is the base clock signal and is provided to the clock
input of D flip-flop 930 and to the anodes of rectifiers 931 and 934. The
output of inverter 929 is designated as "CK" in FIG. 11A and is shown in a
timing graph 1002 of FIG. 12.
As indicated above, the base clock signal at the output of inverter 929 is
coupled to the clock input of flip-flop 930. Additionally, the
complementary output Q is coupled to the data input D. With this
configuration, as known in the digital art, the output Q of flip-flop 930
toggles between logic 0 and logic 1 each time the clock input undergoes a
logic low-to-high transition. The Q output of flip-flop 930 is shown in a
timing diagram 1003 of FIG. 12. The toggle action of the output with
respect to the clock input may be seen by comparing timing graphs 1002 and
1003. The output Q of flip-flop 930 is the complement of the Q output of
flip-flop 930. The complementary output is shown in a timing graph 1004 in
FIG. 12.
The Q output flip-flop 930 is coupled to the anode terminal of rectifier
932, which is a first input of the two-input OR gate formed by rectifiers
931 and 932 and resistors 935 and 936. The second input of this OR gate is
provided by the anode of rectifier 931, which is coupled to the base clock
(output of inverter 929). The output of this OR gate is coupled to
terminal 921 and provides the first clock signal .PHI..sub.1 thereto.
Resistors 935 and 936 are coupled in series to provide the "pull-down
resistor" for the OR gate. One terminal of resistor 935 is coupled to
terminal 921 and the other terminal is coupled to each of the cathode
terminals of rectifiers 931 and 932. One terminal of resistor 936 is
coupled to ground and the other terminal is coupled to terminal 921. As
such, the output of the OR gate is generated across resistor 936 and is
logic high whenever the base clock or the Q output of flip-flop 930 is
high. Each of resistors 935 and 936 preferably has a resistance value of
470 ohms. With these values, the first clock signal .PHI..sub.1 has a
logic high value of approximately 2.5 volts whenever the clock output of
inventer 929 or the Q output of flip-flop 930 is logic high. This is more
fully shown in FIG. 12 by logically OR-ing timing graphs 1002 and 1003 to
obtain the timing graph 1005 for the .PHI..sub.1 clock signal at terminal
921.
In a similar manner, the Q output of flip-flop 930 is coupled to the anode
terminal of rectifier 933, which is a first input of the two-input OR gate
formed by rectifiers 933 and 934 and resistors 937-938. The second input
of this OR gate is provided by the anode of rectifier 934, which is
coupled to the base clock (output of inventer 929). The output of this OR
gate is coupled to terminal 922 and provides the second clock signal
.PHI..sub.2 thereto. Resistors 937 and 938 are coupled in series to
provide the "pull-down resistor" for the OR gate. One terminal of resistor
937 is coupled to terminal 922 and the other terminal is coupled to each
of the cathode terminals of rectifiers 933 and 934. One terminal of
resistor 938 is coupled to ground and the other terminal is coupled to
terminal 922. As such, the output of the OR gate is generated across
resistor 938 and is logic high whenever the base clock or the Q output of
flip-flop 930 is high. Each of resistors 937 and 938 preferably has a
resistance value of 470 ohms. With these values, the second clock signal
.PHI..sub.2 has a logic high value of approximately 2.5 volts whenever the
clock output of inventer 929 of the Q output is logic high. This is more
fully shown in FIG. 12 by logically OR-ing timing graphs 1002 and 1004 to
obtain the timing graph 1006 for the .PHI..sub.2 clock signal.
As indicated previously, each of controllers 950 and 960 comprises a
current mode pulse width modulator, preferably an AS3843 PWM controller
manufactured by Astec Semiconductor. Each of controllers 950 and 960
includes a V.sub.CC pin for receiving a source of power at approximately
12 volts, a ground pin GND for receiving a ground reference potential of
zero volts, and R/C clock pin for receiving its respective clock signal
(.PHI..sub.1 or .PHI..sub.2), a V.sub.FB terminal for receiving a signal
representative of the output voltage of the converter (DAC converter 800),
a COMP compensation terminal for providing a compensation signal to be fed
back to the V.sub.FB pin, an I.sub.S current sense pin for receiving a
voltage related to the output inductor current (or primary current), and
an output pin OUT for providing a duty-cycle signal as a function of the
signals present on the R/C, V.sub.FB, COMP, I.sub.S pins. The R/C clock
pin of controller 950 is coupled to terminal 921 of dual clock generator
920 for receiving first clock signal .PHI..sub.1, and the R/C clock pin of
controller 960 is coupled to terminal 922 for receiving the second clock
signal .PHI..sub.2. Additionally, the V.sub.CC pins for each of the
controllers 950 and 960 is coupled to power terminal 918 of internal
supply 915, and each of the ground pins is coupled to ground terminal 917
of internal supply 915. The OUT pin of controller 950 provides the first
duty-cycle signal at port 911 and the OUT pin of controller 960 provides
the second duty-cycle signal at port 912. The first and second duty cycle
signals are shown in timing graphs 1007 and 1008, respectively, of FIG.
12, and are designated as S.sub.1 and S.sub.2, respectively, in FIGS. 11
and 12. Each of the remaining pins, COMP, V.sub.FB, and I.sub.S, are
discussed in greater detail below with reference to controller 950.
A voltage proportional to the output voltage of DAC converter 800 is
provided to the V.sub.FB pin of controller 950. This voltage is generated
by a voltage divider comprising two resistors 941 and 942 coupled together
at an intermediate node 940. The voltage divider is coupled to port 827 of
control means 820 and intermediate node 940 provides the proportional
voltage to the V.sub.FB pin of controller 950. In a preferred embodiment
of control means 820, each of resistors 941 and 942 has a resistance value
of 2.2 K-ohms. As such, the voltage at intermediate node 940 is
approximately half of the output voltage of DAC converter 800. Thus, when
the value of the output voltage is at its nominal value of five volts, the
voltage provided to pin V.sub.FB is 2.5 volts.
Within controller 950, the voltage at the V.sub.FB pin is compared to an
internally generated voltage of 2.5 volts. This comparison is preferably
performed by a differential amplifier, often called the error amplifier,
having an inverting input coupled to the V.sub.FB pin and a non-inverting
input coupled to a 2.5 volt reference, which serves to represent the
desired output voltage. The output of the error amplifier provides a
signal which is related to the difference between the desired output
voltage and the measured output voltage. Hence, this voltage is called the
error voltage. As the voltage at the V.sub.FB pin decreases from 2.5
volts, the error voltage increases, and vice-versa. The error voltage is
used to govern the pulse width of the signal S.sub.1 at the output OUT
pin. The pulse width increases as this voltage increases and decreases as
this voltage decreases. Thus, the pulse width of the first duty-cycle
signal at port 911 increases as the sensed output voltage at pin V.sub.FB
decreases from the reference of 2.5 volts and decreases as the sensed
output voltage increases. The increase pulse width causes more power to be
delivered to the output, thus increasing the output voltage and thus the
voltage at pin V.sub.FB. This forms a negative feedback loop which brings
the output voltage in regulation to the targeted 5.0 volts.
As is known in the art, such negative feedback control loops often require
some form of compensation, or frequency-dependent damping, to ensure that
the control loop is stable and does not oscillate in an uncontrolled
manner. When the control loop is operated without compensation, the output
voltage will typically oscillate at a natural resonant frequency of the
control loop. This natural resonant frequency is due to the fact that
there is finite delay between the time that the error amplifier causes a
correction in the duty cycle of the PWM output signal and the time the
converter's filter inductor and output capacitor respond to the corrected
duty cycle. In general, the error amplifier can generate an error
correction signal to the pulse width modulation faster than the output
filter and capacitor can react. A compensation network comprising of two
capacitors 951 and 952 and a resistor 953 is connected between the COMP
and V.sub.FB pins of controller 950. The compensation network is, in turn,
coupled between the inverting input of the error amplifier and the output
of the error amplifier. The compensation network feeds back a portion of
the output signal to the inverting input so as to slow the operation of
the error amplifier down to the response time scale of the output filter
and output capacitor. In this way, errant oscillations in the output
voltage are mitigated. Exemplary component values for this compensation
network are: 2.2 nF for capacitor 951, 100 pF for capacitor 952, and 8.3
K-ohms for resistor 953.
The output voltage of the error amplifier controls the pulse width at the
OUT pin in the following manner. The pulse state of the signal S.sub.1 at
the OUT pin is initiated by the rising edge at the R/C clock pin. This is
shown at 1021 in graph 1007 of FIG. 12. To determine when the pulse state
is to end, the error voltage from the error amplifier is compared to the
voltage present on the I.sub.S pin. The I.sub.S voltage is proportional to
the output filter inductor current of converter 100 during the converter's
ON period (i.e., during the pulse state at the OUT pin). In the present
invention, this current is sampled by way of the primary circuit current
I.sub.IN, which is shown at timing graph 860 in FIG. 10. It has a
trapezoidal shape, with the slope of the trapezoid being determined by the
input voltage, the inductance of the output filter inductor 170 of first
converter 100, and the magnetizing inductance of inductor 123 of first
converter 100. This current is preferably sensed by a current transformer
(e.g., transformer 135 in FIG. 9) and coupled to a resistor to convert it
to a voltage, which is then provided to the I.sub.S pin. In operation, the
output error voltage sets a threshold level which the voltage signal at
the I.sub.S pin must reach before controller 950 ends the pulse state of
the signal at the OUT pin (i.e. the first duty-cycle signal). However, if
the I.sub.S voltage does not reach this threshold before the falling edge
of the signal at the R/C clock pin occurs, controller 950 will terminal
the pulsed state at the OUT pin. This limits the maximum duty-cycle of the
signal at the OUT pin to be substantially equal to the duty cycle of the
signal at the R/C clock pin, which is preferably 0.75 (75%). The maximum
duty cycle for the signal S.sub.1 is shown at 1022 in graph 1007 of FIG.
12.
In a preferred embodiment of controller 950, the output error signal is
preferably level-shifted and/or scaled before it is compared to the
I.sub.S voltage so that it varies between a value of zero and 1 V.
Additionally, the I.sub.S signal is generated such that it has a range of
zero to .about.1 V, with the maximum value of .about.1 V corresponding to
the maximum allowable current through output filter inductor 170. In a
preferred embodiment of the present invention, the maximum output current
for inductor 170 is approximately 23 amperes.
Depending upon the load coupled to the converter, the magnitude of the load
current may vary between zero and its maximum value even though the output
voltage is at its nominal value. This means that the voltage at the
I.sub.S pin may vary between near zero volts and .about.1 V even though
the output voltage is at its nominal value. This would appear to be
incompatible with the threshold level set by the error amplifier output.
However, the negative feedback in the control loop automatically adjusts
the threshold level set by the error amplifier to account for changes in
the load current. In other words, the threshold level set by the error
amplifier is a function of both the output voltage and the average load
current.
As known in the art, the comparison of the I.sub.S current with the output
error voltage comprises a current mode control loop. Such a current mode
control has several advantages. As a first advantage, the current control
mode scheme can instantaneously correct for input voltage variations
without having the error amplifier swing through large dynamic ranges.
This is because the slope of the I.sub.S current is related to the input
voltage, decreasing (becoming more shallow) as the input voltage decreases
from the nominal value and increasing (becoming steeper) as the input
voltage increases above the nominal value. This allows the duty-cycle to
vary within one switching cycle of a change in the input voltage. For
example, if the input voltage decreases by a predetermined amount, the
slope will also decrease by a predetermined amount, which in turn will
extend the duration of the pulse. (i.e., increase the duty-cycle). This in
turn increases the amount of power provided to the load within one
switching cycle. As a second advantage, the compensation network can be
simplified over the conventional voltage mode control loops. This allows
for faster response and fewer components.
The voltage signal at the I.sub.S pin of controller 950 is generated in the
following manner. A current proportional to the primary circuit current of
the first converter 100 is coupled to port 828 of control means 820. This
is provided by the current sensing transformer 135 shown in FIG. 9, as
discussed above. This current is typically 1/40th of the current
(I.sub.IN) flowing in the primary circuit. This current is fed to a
resistor 944, which generates a voltage in proportion to the current
provided at port 828. In a preferred embodiment of control means 820,
resistor 944 has a value of 5 ohms and is preferably formed by coupling
two 10 ohm resistors in parallel. The voltage developed across resistor
944 is then coupled to the I.sub.S pin by way of a resistor 945 and a
capacitor 946. Resistor 945 and capacitor 946 filter the voltage developed
across resistor 944 such that any leading edge spikes are removed. Such
leading edge spikes may cause controller 950 to prematurely end the pulse
period of the signal at its OUT pin. In a preferred embodiment, resistor
945 has a value of approximately 200 ohms and capacitor 946 has a value of
approximately 100 pF.
As is known in the current art, current control mode schemes have stability
problems when the duty-cycle exceeds 50%. This instability is
unconditional and occurs regardless of the state of the voltage feedback
loop. It is known that by adding a ramp voltage to the voltage generated
at resistor 944 this instability can be eliminated. With this added ramp
voltage, a current mode controlled converter may operate at duty-cycles
greater than 50%. For generating the added ramp voltage, PWM means 910
comprises a ramp generation means including a capacitor 956, a rectifier
958, and three resistors 955, 957, and 959. Resistor 955 and capacitor 956
are coupled in series, and the series combination is coupled between the
OUT pin of controller 950 and ground. At the beginning of the pulse
period, the voltage at the OUT pin switches from zero volts to
approximately 10 volts. This generates an R/C ramp voltage across
capacitor 956. This ramp is coupled to the I.sub.S terminal by resistor
957. At the end of the pulse period, the voltage signal at the OUT pin
switches from approximately 10 volts to zero volts. Rectifier 958 and
resistor 959 discharge the voltage built up on capacitor 956. Rectifier
958 and resistor 959 are coupled in series, and this series combination is
coupled between the OUT pin and capacitor 956.
During the pulse period, or ON period, the rising voltage at capacitor 956
injects a current into resistor 945 by way of resistor 957. This injected
current causes a voltage drop across resistor 945, which adds to the
voltage generated across resistor 944, as indicated above. In this way,
the compensation ramp voltage is added to the voltage proportional to the
primary circuit current. Preferred values for these components are as
follows: resistor 955, 2.2 K-ohm; capacitor 956, 1 nF; resistor 957, 2.2
K-ohms; and resistor 959, 220 ohms. With these values, the ramp voltage
across capacitor 956 has a time constant of approximately 1.2
microseconds. With the value of 220 ohms for resistor 945, approximately
1/10 th of the ramp voltage at the capacitor 956 is generated across
resistor 945. At a 50% duty-cycle and a full load of 20 A per converter,
resistor 944 provides approximately 0.4 volts to the I.sub.S pin by the
end of the pulse period and resistor 945 provides approximately 0.7 volts
to the I.sub.S pin by the end of the pulse period. As is known in the art,
the slope of the ramp, as provided across resistor 945, must be greater
than 1/2 of the absolute value of the "ripple down slope". The ripple down
slope is the slope of the voltage that would be generate across resistor
944 if the output filter inductor's current during the OFF period could be
coupled to resistor 944 during the OFF period. Since primary switch means
140 of converter 100 is opened during the OFF period, the output filter
inductor's current is prevented from being coupled to resistor 944.
Nonetheless, given the nominal output voltage, the inductance of filter
inductor 170, and the turns ratios of transformers 120 and 135, the ripple
down current may be readily calculated.
The functions of the COMP, V.sub.FB, and I.sub.S pins for second controller
960 are the same as those of the corresponding pins of controller 950.
However, different signals are coupled to the COMP and V.sub.FB pins of
second controller 960. Specifically, the V.sub.FB pin is grounded, which
effectively disables the error amplifier of second controller 960.
Additionally, the COMP pin of first controller 950 is coupled to the COMP
pin of second controller 960, causing the output error signal generated by
controller 950 to be used as the output error signal for controller 960.
This causes the duty cycles generated at the OUT pins of controllers 950
and 960 to be substantially the same since the same threshold level is
used to compare the voltages at the I.sub.S pins of controllers 950 and
960.
The voltage at the I.sub.S pin of controller 960 is generated in the same
manner as that for controller 950. Specifically, a current proportional to
the primary circuit current of second converter 100' is coupled to port
829 of control means 820. This is provided by the current sensing
transformer 135' shown in FIG. 9, as discussed above. This current is
typically 1/40 th of the current (I'.sub.IN) flowing in the primary
circuit. This current is fed to a resistor 947, which generates a voltage
in proportion to the current provided at port 829. Resistor 947 preferably
has the same resistance as resistor 944. The voltage developed across
resistor 947 is coupled to the I.sub.S pin by way of a resistor 948 and a
capacitor 949. Resistor 948 preferably has the same resistance as resistor
945 and capacitor 949 preferably has the same capacitance as capacitor
946.
As with controller 950, a ramp voltage from the OUT pin of controller 960
is added to the I.sub.S pin of controller 960 to ensure stable operation
for duty cycles above 50%. The ramp generation circuit is substantially
the same as that for controller 950 and comprises a capacitor 966, a
rectifier 968, and three resistors 965, 967, and 969. These components
have substantially the same component values as capacitor 956, rectifier
958, and resistors 955, 957, and 959, respectively.
Having discussed the operation of PWM means 910, the operation of first
control means 970 is discussed. As indicated above, first control means
970 is responsive to the first duty-cycle signal S.sub.1 at port 911 and
is for operating primary switch means 140 by way of port 821, first switch
means 130 by way of port 822, and second switch means 150 by way of port
823 in the manner indicated above. The control signals at ports 821-823
are shown in timing graphs 1009, 1010, and 1013 respectively, of FIG. 12.
First control means 970 comprises a first transistor drive circuit for
driving primary switch means 140, a second transistor drive circuit for
driving first switch means 130, and a third transistor drive circuit for
driving second switch means 150.
The first drive circuit comprises a buffer amplifier 972 having an input
and an output, an isolation transformer 974 having a primary winding
coupled to the output of amplifier 972 and a secondary winding coupled to
port 821, a first coupling capacitor 973 coupled between the output of
amplifier 972 and the primary winding of transformer 974, and a second
coupling capacitor 975 coupled between the secondary winding of
transformer 974 and port 821. Buffer amplifier 972 is responsive to the
signal at port 911 and provides the necessary power for driving primary
switch means 140, transformer 974 provides isolation between the different
ground references used by control means 820 and primary switch means 140,
and coupling capacitors 973 and 975 prevent transformer 974 from becoming
saturated. The first drive circuit further comprises a rectifier 976 for
clamping the negative voltage at port 821 and for establishing a "ground"
reference potential for port 821. In operation, amplifier 972 provides an
output of approximately 10 V when signal S.sub.1 is in its pulse state and
an output of approximately 0 V when signal S.sub.1 is in its resting
state. Transformer 974 and capacitors 973 and 975 couple the output of
amplifier 972 to port 821.
The first drive circuit further comprises a delay unit coupled between port
911 and the input of amplifier 972. The delay unit includes a resistor 977
and a rectifier 978, each of which are coupled between port 911 and the
input of amplifier 972 (cathode of rectifier 978 coupled port 911), and a
capacitor 979 coupled between the input of amplifier 972 and ground. This
delay unit, along with a similar delay unit in the second transistor drive
circuit described below, comprises means for ensuring that the control
signals at ports 821 and 822 are not pulsed at the same time, thereby
preventing both switch means 130 and 140 from conducting at the same time.
In response to the leading edge transition of the pulse state for the
first duty cycle signal S.sub.1, resistor 977 and capacitor 979 function
as an RC delay means which delays the propagation of the leading edge of
signal S.sub.1 to the input of amplifier 972. This causes a delay between
the leading edge of signal S.sub.1 and the corresponding leading edge of
the control signal at port 821. This delay is shown at 1025 in graph 1009.
In response to the falling edge transition of signal S.sub.1, rectifier
978 discharges capactor 979 with little delay. As such, relatively little
delay occurs between the falling edge of signal S.sub.1 and the
corresponding falling edge of the control signal at point 821. This is
shown at 1026 in graph 1009 of FIG. 12.
The second drive circuit comprises an inverting buffer amplifier 982 having
an input and an output, an isolation transformer 984 having a primary
winding coupled to the output of amplifier 982 and a secondary winding
coupled to port 822, a third coupling capacitor 983 coupled between the
output of inverting amplifier 982 and the primary winding of transformer
984, and a fourth coupling capacitor 985 coupled between the secondary
winding of transformer 984 and port 822. Inverting amplifier 982 is
responsive to the signal S.sub.1 at port 911 and provides the necessary
power for driving first switch means 130, transformer 984 provides
isolation between the different ground references used by control means
820 and first switch means 130, and coupling capacitors 983 and 985
prevent transformer 984 from becoming saturated. The second drive circuit
further comprises a rectifier 986 for clamping the negative voltage at
port 822 and for establishing a "ground" reference potential for port 822.
In operation, inverting amplifier 982 provides an output of approximately
0 V when signal S.sub.1 is in its pulse state and an output of
approximately 10 V when signal S.sub.1 is in its resting state.
Transformer 984 and capacitors 983 and 985 couple the output of amplifier
982 to port 822.
The second drive circuit further comprises a delay unit coupled between
port 911 and the input of inverting amplifier 982. The delay unit includes
a resistor 987 and a rectifier 988, each of which are coupled between port
911 and the input of amplifier 982 (anode of rectifier 988 coupled port
911), and a capacitor 989 coupled between the input of amplifier 982 and
ground. The orientation of rectifier 988 is the reverse of the orientation
of rectifier 978. In response to the rising edge transition of the pulse
state for the first duty cycle signal S.sub.1, rectifier 988 charges up
capacitor 989 relatively quickly. As such, relatively little delay occurs
between the rising edge of signal S.sub.1 and the corresponding falling
edge of the control signal at port 822. This is shown at 1028 in graph
1010 of FIG. 12. In response to the falling edge transition of signal
S.sub.1, rectifier 988 is non-conducting and resistor 987 and capacitor
989 functions as an RC delay means which delays the propagation of the
falling edge of signal S.sub.1 to the input of inverting amplifier 982.
This causes a delay between the falling edge of signal S.sub.1 and the
corresponding leading edge of the control signal at port 822. This delay
is shown at 1029 in graph 1010.
This delay unit, along the delay unit in the first transistor drive
circuit, comprises means for ensuring that the control signals at ports
821 and 822 are not pulsed at the same time, thereby preventing both
switch means 130 and 140 from conducting at the same time. This may be
seen by comparing graphs 1009 and 1010 shown in FIG. 12 and by noting the
locations of the time delays at 1025 and 1029.
The third drive circuit comprises a buffer amplifier 992 having an input
and an output, an isolation transformer 994 having a primary winding
coupled to the output of amplifier 992 and a secondary winding coupled to
port 823, a fifth coupling capacitor 993 coupled between the output of
amplifier 992 and the primary winding of transformer 994, and a sixth
coupling capacitor 995 coupled between the secondary winding of
transformer 994 and port 823. Buffer amplifier 992 is responsive to the
signal at the output of buffer amplifier 972, which drives primary switch
means 140, and provides the necessary power for driving second switch
means 150. Transformer 994 provides isolation between the different ground
references used by control means 820 and second switch means 150, and
coupling capacitors 993 and 995 prevent transformer 994 from becoming
saturated. The third drive circuit further comprises a rectifier 996 for
clamping the negative voltage at port 823 and for establishing a "ground"
reference potential for port 823. In operation, amplifier 992 provides an
output of approximately 10 V when the output signal of amplifier 972 is in
its pulse state (.about.+10 V) and an output of approximately 0 V when the
output signal of amplifier 972 is in its resting state (.about.0 V).
Transformer 994 and capacitors 993 and 995 couple the output of amplifier
992 to port 823.
The third drive circuit further comprises a delay unit coupled between the
output of buffer amplifier 972 and the input of amplifier 992. The delay
unit includes a resistor 997 and a rectifier 998, each of which are
coupled between the output of amplifier 972 and the input of amplifier 992
(cathode of rectifier 978 coupled to the output of amplifier 972), and a
capacitor 991 coupled between the input of amplifier 992 and ground. This
delay unit causes second switch means 150, by way of the control signal at
port 823, to close a time duration after primary switch means 140 closes,
as directed by the control signal at port 921, and to open a short time
duration after primary switch means 140 opens. In response to the leading
edge transition of the pulse state at the output of amplifier 972,
resistor 997 and capacitor 991 function as an RC delay means which delays
the propagation of the leading edge of the output of amplifier 972 to the
input of amplifier 992. This causes a delay between the leading edge of
control signal at port 821 and the corresponding leading edge of the
control signal at port 823. This delay is shown at 1030 in graph 1013. In
response to the falling edge transition of the pulse state at the output
of amplifier 972, rectifier 998 discharges capacitor 991 with little
delay. As such, relatively little delay occurs between the falling edge of
the control signal at port 821 and the corresponding falling edge of the
control signal at port 823. This is shown at 1031 in graph 1013 of FIG.
12.
In a similar manner, second control means 999 shown in FIG. 11B is
responsive to the second duty-cycle signal S.sub.2 at port 912 and is for
operating primary switch means 140', first switch means 130', and second
switch means 150' in the manner indicated above. Second control means 999
provides a control signal for primary switch means 140' at control port
824, a control signal for first switch means 130' at control port 825, and
an control signal for second switch means 150' at control port 826. The
control signals at ports 824-826 are shown in timing graphs 1011, 1012,
and 1014 respectively, of FIG. 12. The structure and operation of second
control means 999 is substantially identical to those of first control
means 970. Therefore, reference is made to the above discussion of first
control means 970 for a detailed description of second control means 999.
A further object of the present invention is to provide a switch control
means for second switch means 150 of the present invention which is
autonomous from the control circuitry for switch means 130 and 140, i.e.,
autonomous from PWM means 910 and buffer 972. The autonomous feature of
the switch control means for second switch means 150 provides greater
manufacturing flexibility and circuit robustness. Such an exemplary
autonomous switch control means is shown at 730 in FIG. 11C and may be
used in place of the third transistor drive circuit comprising elements
991-998 shown in FIG. 11B. Switch control means 730 is shown in the
context of a partial block diagram 700 of forward converter 100, which is
shown in FIG. 4, along with switching transistor 752, diode 754, and
blocking rectifier 756.
The present invention recognizes that a control signal for controlling
second switch means 150 may be derived from the voltage across secondary
winding 124 in the following manner. Referring back to timing diagram 300
shown in FIG. 6, it may be seen that for switching sequence 230 (of FIG.
5), the voltage waveform V.sub.s for secondary winding 124 crosses zero
volts, as indicated at a point 314 in FIG. 6, at substantially the same
time second switch means 150 is closed. The zero crossing occurs as
V.sub.s changes from a positive value to a negative value and is referred
to as a negative transition due to the negative slope of the V.sub.s
waveform. It may be further seen from timing diagram 300 that second
switch means 150 thereafter opens a time duration after the voltage
waveform V.sub.s for secondary winding 124 crosses zero volts, indicated
at a point 315 in FIG. 6, as V.sub.s changes from a negative value to a
positive value. This is referred to as a positive transition due to the
positive slope of the V.sub.s waveform.
In the case of switching sequence 230, it may then be appreciated that
second switch means 150 may be controlled in response to the voltage
V.sub.s of secondary winding 124 by detecting the transitions of the
V.sub.s waveform with respect to a predetermined voltage value, e.g., zero
volts, and by directing second switch means 150 to open a short time
(short time duration) after voltage V.sub.s makes a negative transition
through the predetermined voltage value and to close a somewhat longer
time (longer time duration) after voltage V.sub.s makes a positive
transition through the predetermined voltage value. The length of the
second duration may be chosen (i.e., made large enough) such that second
switch means 150 is closed after primary switch means 140 closes for a
wide range of duty-cycles and load conditions. In this manner, a switch
control means may be constructed for second switch control means 150 which
may be autonomous from the switch control means for primary switch means
140 and first switch means 130. Such an autonomous switch control means is
provided by switch control means 730 shown in FIG. 11C.
Autonomous switch control means 730 comprises a series combination of a
resistor 732 and a Zener diode 734 which is coupled in parallel across
secondary winding 124. The cathode terminal of Zener diode 734 is coupled
to a node 738 and the anode terminal of Zener diode 734 is connected to
the negative terminal of secondary winding 124, which serves as a
secondary-side ground. Resistor 732 is connected between the positive
terminal of secondary winding 124 and node 738. Additionally, a rectifier
736 is coupled in parallel with Zener diode 734 with its cathode terminal
coupled to the secondary-side ground.
In operation, a positive voltage of, for example, 4.8 V is generated at
node 738 by Zener diode 734 when a positive potential appears across
secondary winding 124 to direct current towards load 174. When a negative
voltage appears across secondary winding 124, a negative voltage of
approximately -0.6 V is generated at node 734 by rectifier 736. As such,
it may be appreciated that resistor 732, diode 734, and rectifier 736
comprise detection means for detecting the voltage across secondary
winding 124 and for generating a signal at node 738 having a first state
(4.8 V, logic high) when the voltage across secondary winding 124 is
positive and a second state (-0.6 V, logic low) when the voltage across
secondary winding 124 is negative.
Autonomous switch control means 730 further comprises a delay generation
means responsive to the signal at node 738 for generating a switch control
signal at a node 746 to control second switch means 150. As discussed
below in greater detail, the control signal at node 746 has a first state
(logic-high) for directing second switch means 150 to close and a second
state (logic low) for directing second switch means 150 to open. The delay
generation means comprises a first inverter 740 having an input coupled to
node 738 and an output, a second inverter 745 having an input coupled to
the output of first inverter 740 and an output terminal coupled to node
746 for providing the control signal. Inverters 740 and 745 are powered by
the potential difference between a conventional digital V.sub.CC supply
(for example +5 V) and the secondary-side ground. As such, the outputs of
inverters 740 and 745 have a logic low state at around zero volts and a
logic high state at around V.sub.CC (+5) volts. In addition, inverter 745
preferably comprises a Schmitt trigger input, as described below in
greater detail.
The delay generation means of control means 730 further comprises a delay
means having a rectifier 741, a resistor 742 and a capacitor 744 for
introducing first and second predetermined time delays into the signal
propagation between first inverter 740 and second inverter 745.
Specifically, the delay means introduces a first predetermined signal
propagation delay between first inverter 740 and second inverter 745 when
the output of inverter 740 makes a transition from a logic-low level to a
logic-high level. Additionally, the delay means introduces a second
predetermined signal propagation delay between first inverter 740 and
second inverter 745 when the output of inverter 740 makes a transition
from a logic-high level to a logic-low level. In the delay means, the
anode terminal of rectifier 741 is connected to the output of inverter 740
and the cathode terminal of rectifier 741 is connected to the input of
inverter 745 at a node 743. Capacitor 744 is coupled between the input of
inverter 745 and ground and resistor 742 is also coupled between the input
of inverter 745 and ground.
When the output of inverter 740 makes a transition from a logic-low level
to a logic-high level, rectifier 741 conducts and current from inverter
740 charges capacitor 744 to raise the voltage at node 743. The component
values of rectifier 741, resistor 742, and capacitor 744 are chosen in a
manner well known in the art such that the voltage at node 743 rises to a
logic-high value in a time duration which is relatively short with respect
to the switching cycle time. This provides the above first predetermined
signal propagation delay. When the output of inverter 740 makes a
transition from a logic-high level to a logic-low level, rectifier 741
blocks current from flowing from capacitor 744 into inverter 740 and, as
such, the charge on capacitor 744 supporting the voltage on node 743 is
discharged to ground by resistor 742. The component values of resistor 742
and capacitor 744 are chosen in a manner well known in the art such that
the voltage at node 743 falls to a logic-low level in a time duration
which is relatively longer than the rise time duration for the voltage at
node 743. This provides the above second predetermined signal propagation
delay.
Given the slow rise time at node 743, it may be appreciated that the use of
a Schmitt trigger at the intut of inverter 745 (i.e., a Schmitt triggered
inverter) would improve the output switching characteristics of inverter
745. It may also be appreciated that means other than rectifier 741,
resistor 742, capacitor 744, and the particular arrangement of these
components may be used to provide the first and second predetermined
propagation delays.
The control signal at node 746 tracks the detected signal at node 738 and
is delayed with respect to the signal at node 738 by one of two
predetermined time durations, the time duration being selected by the
direction of the logic transition (positive transition and negative
transition). As a result, the control signal at node 746 enters its first
state (logic-high level) a first predetermined time duration after the
signal at node 738 enters its first state (logic-high level), the first
predetermined time duration being related to the second (longer) signal
propagation delay generated by resistor 742 discharging capacitor 744.
Further, the control signal at node 746 enters its second state (logic-low
level) a second predetermined time duration after the signal at node 738
enters its second state (logic-low level), the second predetermined time
duration being related to the first (shorter) signal propagation delay
generated by the charging of capacitor 744 through rectifier 741.
The control signal generated at node 746 in this way is provided to second
switch means 150 via a coupling means including a coupling transformer
780. Coupling transformer 780 includes a first winding 782 and a second
winding 784. First winding 782 has a first terminal coupled to node 746
for receiving the switch control signal and a second terminal coupled to
the secondary-side ground. Second winding 784 has a first terminal coupled
to the gate of transistor 752 and a second terminal coupled to the source
of transistor 752. The magnetic reference convention for transformer 780
is such that a reference dot appears at the first terminals of windings
782 and 784. As is well known to the transformer coupling art, a resistor
786 may be included in series with second winding 784 to damp any
oscillations that may occur by any resonance formed by the gate
capacitance of transistor 752 and the leakage inductance of second winding
784.
The configuration of transformer 780 in this manner provides means for
translating the potential voltage generated at node 746 to a potential
difference generated across the gate and source terminals of transistor
752. As a result, transistor 752 conducts current between its drain and
source terminals when the control signal at node 746 is in its first state
(logic-high level) since the positive voltage at node 746 is translated by
transformer 780 to provide a positive gate-to-source voltage for
transistor 752. Additionally, transistor 752 blocks current between its
drain and source terminals when the control signal at node 746 is in its
second state (logic-low level) since the near zero voltage at node 746 is
translated by transformer 780 to provide a near zero gate-to-source
voltage for transistor 752. It may thus be appreciated that transformer
780 comprises means for coupling the control signal at node 746 to
transistor 752 of second switch means 150.
Thus, it may be appreciated that, in place of the third drive circuit, the
autonomous switch control means 730 described above may be used to control
second switch means 150 and that a corresponding autonomous switch control
means 730' may also be used to control second switch means 150'. In such a
case, each of the autonomous switch control means 730 and 730' are
responsive to the voltage across the transformer secondary windings of
converters 100 and 100', respectively, as described above.
While the invention has been described in connection with what is presently
considered to be the most practical and preferred embodiments, it is to be
understood that the present invention is not limited to the disclosed
embodiments but, on the contrary is intended to cover various
modifications and equivalent arrangements included within the scope of the
appended claims.
Top